Re: [time-nuts] Cable length calibration

[Mike Cook]

> Le 29 juin 2016 à 22:18, Poul-Henning Kamp  a écrit :
> 
> 
> In message <20160629192850.19c29406...@ip-64-139-1-69.sjc.megapath.net>, Hal 
> Mu
> rray writes:
> 
>>> At one point they were looking into making a GPS time receiver where the
>>> cable length calibration would be built-in. 
>> 
>> How would you do that?
> 
> TDR ?
> 
> If it wasn't behind a choke, the inrush current to the antenna
> preamp power filtering capacitor could be measured, but the choke
> ruins that.
> 
> The trouble is how to do it without frying the antenna preamp...
> 
> 
> Seriously...
> 
> GPS antennas and receivers are cheap, I would just use two GPS antennas
> with a known difference in cable-length.
> 
Sounds simple, but even after a days reflection I don’t see how you 
find the complete path delay. You would get the cable delay (OPs concern) 
provided they were the same antenna/cable type combinations, but not delay 
induced by the antenna electronics.  From another post that delay seems to be 
non-negligable. I find it curious that antenna manafacturers don’t seem to give 
this parameter. I looked at some datasheets on Trimble and Leica sites but they 
don’t have it. 
As for me, I measure cable delay by injecting a 1PPS into it through a T and 
for the RG174 attached to the patch antennas, just sacrificed one by cutting 
the head off and measuring that. 

> -- 
> Poul-Henning Kamp   | UNIX since Zilog Zeus 3.20
> p...@freebsd.org | TCP/IP since RFC 956
> FreeBSD committer   | BSD since 4.3-tahoe
> Never attribute to malice what can adequately be explained by incompetence.
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"The power of accurate observation is commonly called cynicism by those who 
have not got it. »
George Bernard Shaw

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[time-nuts] DMTD downmixer question

[Stéphane Rey]

Hi there,

I will receive my SR620 soon and want of course to use it as well for 
stability measurement using the DMTD method.


I've read many things on how to design the downmixer. There will be a 
DDS or low noise generator as LO, the two mixers, and the squarer. There 
are apparently many ways to do the squarer. Some of the ways I've seen 
are using fast comparator, logic gate, fast amplifiers...
Finally is there a way which looks better than the others ? I was 
hesitating between a fast comparator and an ECL logic gate for instance.


Thanks & cheers
Stephane

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Re: [time-nuts] Cable length calibration

[Bob Camp]

Hi


> On Jul 1, 2016, at 3:40 AM, Mike Cook  wrote:
> 
> 
>> Le 29 juin 2016 à 22:18, Poul-Henning Kamp  a écrit :
>> 
>> 
>> In message <20160629192850.19c29406...@ip-64-139-1-69.sjc.megapath.net>, Hal 
>> Mu
>> rray writes:
>> 
 At one point they were looking into making a GPS time receiver where the
 cable length calibration would be built-in. 
>>> 
>>> How would you do that?
>> 
>> TDR ?
>> 
>> If it wasn't behind a choke, the inrush current to the antenna
>> preamp power filtering capacitor could be measured, but the choke
>> ruins that.
>> 
>> The trouble is how to do it without frying the antenna preamp...
>> 
>> 
>> Seriously...
>> 
>> GPS antennas and receivers are cheap, I would just use two GPS antennas
>> with a known difference in cable-length.
>> 
>   Sounds simple, but even after a days reflection I don’t see how you 
> find the complete path delay. You would get the cable delay (OPs concern) 
> provided they were the same antenna/cable type combinations, but not delay 
> induced by the antenna electronics.  From another post that delay seems to be 
> non-negligable. I find it curious that antenna manafacturers don’t seem to 
> give this parameter. I looked at some datasheets on Trimble and Leica sites 
> but they don’t have it. 
> As for me, I measure cable delay by injecting a 1PPS into it through a T and 
> for the RG174 attached to the patch antennas, just sacrificed one by cutting 
> the head off and measuring that. 

If you dig into the papers on calibrating the delay of the whole antenna, there 
are a bunch of things they run into. I suspect 
the biggest one is quite simple: nobody but TimeNuts care. The survey guys are 
happy once they know the phase center. The
NIST category guys are going to calibrate it anyway. The cell phone outfits 
never did any real GPS time calibration, they did 
it all on the other end of the system. Way to many cables in a cell site to 
measure them all ….

Bob

> 
>> -- 
>> Poul-Henning Kamp   | UNIX since Zilog Zeus 3.20
>> p...@freebsd.org | TCP/IP since RFC 956
>> FreeBSD committer   | BSD since 4.3-tahoe
>> Never attribute to malice what can adequately be explained by incompetence.
>> ___
>> time-nuts mailing list -- time-nuts@febo.com
>> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
>> and follow the instructions there.
> 
> "The power of accurate observation is commonly called cynicism by those who 
> have not got it. »
> George Bernard Shaw
> 
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Re: [time-nuts] 74HCT9046A Max. Operating Frequency

[Daniel Mendes]

Bringing this thread back from death.. a Few days ago  I decided to open 
a case with NXP to find the 9397 750 00078 Application Note. Not only 
they sent me a 76 pages app note, but also a program to help design with 
it. Any site were I can upload them so everybody can take a look?


Daniel


Em 30/04/2014 00:01, Alexander Pummer escreveu:

Him Daniel,
the chip all 74HC4046, HC7046 and the HC9046 are well designed and 
working fine for the application for which they were designed for, but 
of course if you trying to used for something else you may run into 
problems. Roland best describes the funtion of the chip in details 
with many 
examples:http://books.google.com/books/about/Phase_Locked_Loops_6_e_Design_Simulation.html?id=WjNy3RX9xcAC

73
Alex

On 4/29/2014 7:38 PM, Daniel Mendes wrote:


Hi... can you share you routine for designing with this chip? I tried 
using it sometime ago but the results didn´t agree much with what I 
designed, so i gave up (for now... but i´ll return to it :)


Also in the datasheet it says:

13.3 Further information
For an extensive description and application example please refer to 
Application note ordering number 9397 750 00078.


I never found this application note despites all my google-fu. Anyone 
got it?


Daniel


Em 29/04/2014 10:23, sg sg escreveu:
Running my PLL design routine again for 48 kHz I realize that this 
is in fact very advantageous--it greatly reduces the required 
capacitor size in the loop filter. So dividers are clearly the way 
to go.


Samuel
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Re: [time-nuts] Switching transistors, current sources, nonidealties and noise

[Attila Kinali]

Moin,

Thanks everyone for the answers!

On Mon, 20 Jun 2016 01:45:24 -0400
Charles Steinmetz  wrote:

> The transation frequency of the current source transistor is part of the 
> cause, but the primary cause is generally the capacitance of the CS 
> output node to ground.  Some designers put an inductor in series with 
> the output, but I have never found this to be very effective [except in 
> poorly-designed simulations] due to the self-capacitance of the 
> inductor.  Much better, IME, is to add a cascode device to the current 
> source.  (See attached images.)  This has the added benefit of 
> increasing the output resistance.  This increase can be very substantial 
> (several orders of magnitude) if you use a FET cascode device as shown.

I simulated a couple of circuits, with very different results.
First thing that struck me was, that it is neigh impossible to
make cascode circuits stable when using RF transistors. And even
if I managed to do that, small changes in resistor values would
imediatly make it oscillate again, or degrade performance severely.
Same goes for using Darlington circuits (which I tried in order to
minimize the effects of beta variation).

The best results I got was with the attached circuit. Ie using
a classical opamp based npn current source, but using an emitter
follower between transistor and opamp in order to enhance high
frequency (aka transient) performance. R29 is there to load Q7
and to prevent it from going into saturation. R30 is needed for
stabilizing the circuit (I do not exactly understand what the
mechanics of the oscillations are, when R30 is removed, if someone
knows, please tell me). The voltage divider R30/R31 helps to keep
the opamp output away from the lower power rail. If stability is still
an issue, a 5-10pF capacitor should be added from the output of the
opamp to the inverting input (degrades frequency response below 1MHz slightly).

The simulation output shows the current through the (zero) voltage source
at the tail of the differential pair. The I(V11) curve is the circuit as
shown and the I(V7) curve is the same circuit with the two BFU520 replaced
by 2N3904. As can be seen, the transient of the 2N3904 is several times
larger than the one of the BFU520 and lasts for about three times as long.

I have not done any analysis of the temperature stability, yet.
My guess would be that is dominated by the input offset voltage
temperature coefficient of the opamp. But I have no calculations
to prove it.
Noise analysis would be interesting, but I doubt there is enough
data available to actually get some meaningfull results out of it.
 
> > Why do people use general purpose transistors in these places, even
> > though RF transistors definitly improve switching behaviour?
> > I dimply remember that someone said/wrote once, that RF transistors have
> > a higher noise. But if I look at the datasheet, the quoted noise figure
> > for the BFU520 is <1.6dB while the noise figure of the 2N3904 is 2dB best 
> > case.
> 
> I, for one, have said this, but you are not remembering the whole point. 
>   RF transistors are generally considerably noisier AT BASEBAND than GP 
> transistors, both because their geometries are inherently noisier and 
> because they have *much* higher flicker noise corner frequencies 
> (usually 10kHz to some MHz for RF transistors, compared to 10Hz-1kHz for 
> GP transistors).  One might think that this would not matter at RF, but 
> the flicker noise modulates the bias of the transistor (and sometimes 
> other circuit elements), leading to both simple noise modulation as well 
> as phase modulation.  RF transistors are not specified for noise at 
> baseband.

Hmm.. if the flicker noise corner frequency would be in the few 10kHz to
100kHz range, then I would not be worried. The opamp's control loop
should "kill" anything below ~100kHz and dampen quite a bit up to 1-2MHz.
I would even suspect that in the <10kHz range, the noise of the opamp would
dominate the noise of the transistors.

> I modeled the ...01a circuit using a BFR90A BJT as the cascode device, 
> and the simulation showed that the current spikes were reduced by about 
> 50%.  However, my experience tells me that this would not hold in 
> practice.  

Do you know where the discrepancy between simulation and reality comes from?


Attila Kinali

-- 
It is upon moral qualities that a society is ultimately founded. All 
the prosperity and technological sophistication in the world is of no 
use without that foundation.
 -- Miss Matheson, The Diamond Age, Neil Stephenson
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Re: [time-nuts] DMTD downmixer question

[Bruce Griffiths]

The optimum sine to square converter is embodied in the Collins style limiter 
approach consisting of a cascade of limiting amplifiers each with suitable 
individual gain and individual bandwidth, the gain and bandwidth increasing for 
each successive stage until the output slew rate is sufficient to drive a 
comparator without the comparator contributing significant additional jitter.
 Other approaches invariably produce greater output jitter.details here:
Bruce's: Zero Crossing Detectors - KO4BB

|   |
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| Bruce's: Zero Crossing Detectors - KO4BBHelp keep this site free: (More Info) 
   New: The Thunderbolt Monitor Kit is now available! Bruce's: Zero 
Crossing Detectors  |
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|  |
|   |


Bruce

  From: Stéphane Rey 
 To: 'Discussion of precise time and frequency measurement' 
 
 Sent: Friday, 1 July 2016 8:45 PM
 Subject: [time-nuts] DMTD downmixer question
   
Hi there,

I will receive my SR620 soon and want of course to use it as well for 
stability measurement using the DMTD method.

I've read many things on how to design the downmixer. There will be a 
DDS or low noise generator as LO, the two mixers, and the squarer. There 
are apparently many ways to do the squarer. Some of the ways I've seen 
are using fast comparator, logic gate, fast amplifiers...
Finally is there a way which looks better than the others ? I was 
hesitating between a fast comparator and an ECL logic gate for instance.

Thanks & cheers
Stephane

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Re: [time-nuts] Cable length calibration

[Brooke Clarke]

Hi Mike:

For quite a while I was heavily into "chirp" transmissions.  These are HF ionosphere radio transmissions that sweep from 
2 to 30 MHz at 100 kHz/sec.
In order to "tune" the radio to a specific station (you can not tune by frequency) you need to know the start time 
schedule for that specific station (time nuts content).

When GPS became popular the transmitters switched to GPS.
http://www.prc68.com/I/RCS-5A.shtml

You can use a pulse of RF to calibrate the time delay through your HF receiver to get a more accurate time of reception 
value.  That helps because with a GPS synchronized transmitter you can determine it's great circle distance from  you.  
Under some conditions you can see a transmission going around the Earth 2 or 3 times.


In a similar way if you used a pair non amplified versions of a GPS antenna back to back you could determine the time 
delay of the pair and then divide by two.


--
Have Fun,

Brooke Clarke
http://www.PRC68.com
http://www.end2partygovernment.com/2012Issues.html
The lesser of evils is still evil.

 Original Message 

Le 29 juin 2016 à 22:18, Poul-Henning Kamp  a écrit :


In message <20160629192850.19c29406...@ip-64-139-1-69.sjc.megapath.net>, Hal Mu
rray writes:


At one point they were looking into making a GPS time receiver where the
cable length calibration would be built-in.

How would you do that?

TDR ?

If it wasn't behind a choke, the inrush current to the antenna
preamp power filtering capacitor could be measured, but the choke
ruins that.

The trouble is how to do it without frying the antenna preamp...


Seriously...

GPS antennas and receivers are cheap, I would just use two GPS antennas
with a known difference in cable-length.


Sounds simple, but even after a days reflection I don’t see how you 
find the complete path delay. You would get the cable delay (OPs concern) 
provided they were the same antenna/cable type combinations, but not delay 
induced by the antenna electronics.  From another post that delay seems to be 
non-negligable. I find it curious that antenna manafacturers don’t seem to give 
this parameter. I looked at some datasheets on Trimble and Leica sites but they 
don’t have it.
As for me, I measure cable delay by injecting a 1PPS into it through a T and 
for the RG174 attached to the patch antennas, just sacrificed one by cutting 
the head off and measuring that.


--
Poul-Henning Kamp   | UNIX since Zilog Zeus 3.20
p...@freebsd.org | TCP/IP since RFC 956
FreeBSD committer   | BSD since 4.3-tahoe
Never attribute to malice what can adequately be explained by incompetence.
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have not got it. »
George Bernard Shaw

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Re: [time-nuts] Cable length calibration

[jimlux]

On 7/1/16 9:04 AM, Brooke Clarke wrote:

Hi Mike:

For quite a while I was heavily into "chirp" transmissions.  These are
HF ionosphere radio transmissions that sweep from 2 to 30 MHz at 100
kHz/sec.
In order to "tune" the radio to a specific station (you can not tune by
frequency) you need to know the start time schedule for that specific
station (time nuts content).
When GPS became popular the transmitters switched to GPS.
http://www.prc68.com/I/RCS-5A.shtml



I'm building a satellite (actually 2 of them) that is, among other 
things, designed to receive these transmissions.  It turns out that 
accurately measuring the "propagation delay" through the receiver (as in 
from "EM wavefront" to "time stamped samples in the output stream" is 
non trivial.


There's some phase shift/time delay from the physical interaction with 
the antenna and the load impedance presented by the LNA.  Then there's 
the filters and amplifiers in the analog chain. Finally, there's the ADC 
sampling (pipeline delay between voltage at sampling instant to when 
bits appear at the output) delay, and the various delays through the 
digital signal processing (which is fortunately deterministic, but 
non-trivial to actually "measure")


Fortunately, I only claim 10 microsecond timing accuracy so the 100 
kHz/second chirp means that the downconverted stream might have a 
frequency error of 1 Hz.  (that is, if I tell the receiver that the 
chirp starts at 12:34:56.001, and I actually start the ramp at 
12:34:56.00101, I'll see a 1 Hz error in frequency.. if I have a several 
kHz output bandwidth, it will still be in there.


Given that ionospheric delays and propagation delays are substantially 
longer than 10 microseconds, this isn't an issue. 1km is 3 microseconds, 
1000km is 3 milliseconds (or 300 Hz).




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Re: [time-nuts] Switching transistors, current sources, nonidealties and noise

[Bob Camp]

Hi


> On Jul 1, 2016, at 7:56 AM, Attila Kinali  wrote:
> 
> Moin,
> 
> Thanks everyone for the answers!
> 
> On Mon, 20 Jun 2016 01:45:24 -0400
> Charles Steinmetz  wrote:
> 
>> The transation frequency of the current source transistor is part of the 
>> cause, but the primary cause is generally the capacitance of the CS 
>> output node to ground.  Some designers put an inductor in series with 
>> the output, but I have never found this to be very effective [except in 
>> poorly-designed simulations] due to the self-capacitance of the 
>> inductor.  Much better, IME, is to add a cascode device to the current 
>> source.  (See attached images.)  This has the added benefit of 
>> increasing the output resistance.  This increase can be very substantial 
>> (several orders of magnitude) if you use a FET cascode device as shown.
> 
> I simulated a couple of circuits, with very different results.
> First thing that struck me was, that it is neigh impossible to
> make cascode circuits stable when using RF transistors.

Real cascode circuits can be built with RF transistors. They also can be 
simulated.
Simulating them with the “standard” models is a PIA. The issue is that the 
inductance
of the package is not de-embedded from the test “socket” as carefully as it 
might be. 
There is also the somewhat non-intuitive need to stick a low value resistor in 
the base.
Done properly, they are very reproducible and reasonably insensitive to load. 

Bob

> And even
> if I managed to do that, small changes in resistor values would
> imediatly make it oscillate again, or degrade performance severely.
> Same goes for using Darlington circuits (which I tried in order to
> minimize the effects of beta variation).
> 
> The best results I got was with the attached circuit. Ie using
> a classical opamp based npn current source, but using an emitter
> follower between transistor and opamp in order to enhance high
> frequency (aka transient) performance. R29 is there to load Q7
> and to prevent it from going into saturation. R30 is needed for
> stabilizing the circuit (I do not exactly understand what the
> mechanics of the oscillations are, when R30 is removed, if someone
> knows, please tell me). The voltage divider R30/R31 helps to keep
> the opamp output away from the lower power rail. If stability is still
> an issue, a 5-10pF capacitor should be added from the output of the
> opamp to the inverting input (degrades frequency response below 1MHz 
> slightly).
> 
> The simulation output shows the current through the (zero) voltage source
> at the tail of the differential pair. The I(V11) curve is the circuit as
> shown and the I(V7) curve is the same circuit with the two BFU520 replaced
> by 2N3904. As can be seen, the transient of the 2N3904 is several times
> larger than the one of the BFU520 and lasts for about three times as long.
> 
> I have not done any analysis of the temperature stability, yet.
> My guess would be that is dominated by the input offset voltage
> temperature coefficient of the opamp. But I have no calculations
> to prove it.
> Noise analysis would be interesting, but I doubt there is enough
> data available to actually get some meaningfull results out of it.
> 
>>> Why do people use general purpose transistors in these places, even
>>> though RF transistors definitly improve switching behaviour?
>>> I dimply remember that someone said/wrote once, that RF transistors have
>>> a higher noise. But if I look at the datasheet, the quoted noise figure
>>> for the BFU520 is <1.6dB while the noise figure of the 2N3904 is 2dB best 
>>> case.
>> 
>> I, for one, have said this, but you are not remembering the whole point. 
>>  RF transistors are generally considerably noisier AT BASEBAND than GP 
>> transistors, both because their geometries are inherently noisier and 
>> because they have *much* higher flicker noise corner frequencies 
>> (usually 10kHz to some MHz for RF transistors, compared to 10Hz-1kHz for 
>> GP transistors).  One might think that this would not matter at RF, but 
>> the flicker noise modulates the bias of the transistor (and sometimes 
>> other circuit elements), leading to both simple noise modulation as well 
>> as phase modulation.  RF transistors are not specified for noise at 
>> baseband.
> 
> Hmm.. if the flicker noise corner frequency would be in the few 10kHz to
> 100kHz range, then I would not be worried. The opamp's control loop
> should "kill" anything below ~100kHz and dampen quite a bit up to 1-2MHz.
> I would even suspect that in the <10kHz range, the noise of the opamp would
> dominate the noise of the transistors.
> 
>> I modeled the ...01a circuit using a BFR90A BJT as the cascode device, 
>> and the simulation showed that the current spikes were reduced by about 
>> 50%.  However, my experience tells me that this would not hold in 
>> practice.  
> 
> Do you know where the discrepancy between simulation and reality comes from?
> 
> 

Re: [time-nuts] Switching transistors, current sources, nonidealties and noise

[Hal Murray]

kb...@n1k.org said:
> There is also the somewhat non-intuitive need to stick a low value resistor
> in the base. Done properly, they are very reproducible and reasonably
> insensitive to load.  

Is that required for real circuits or just for the simulations?


-- 
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Re: [time-nuts] Switching transistors, current sources, nonidealties and noise

[Scott Stobbe]

There are a plethora of ways to build up a current source. The nice thing
about spice is you can start with a generalized model to see which way you
need to go. For a bipolar current source (sampling current at the emitter)
you are going to achieve a maximum output resistance of beta*ro with an
active servo element as you have included, or with heavy emitter
degeneration. For a basic discrete bjt like a 2n3904, a you will also
include 3-4 pF of output capacitance. For a 20 mA bipolar current source
you are looking an output resistance of 500 kOhm. You can try an ideal 20
mA current source shunted with a 500 kOhm resistor and 4 pF capacitor.

If you need higher output resistance you will have to move to a FET based
approach. If you need less than 1 pF of output capacitance you will need a
better transistor and care in how you physically construct your circuit.

On Fri, Jul 1, 2016 at 2:54 PM, Hal Murray  wrote:

>
> kb...@n1k.org said:
> > There is also the somewhat non-intuitive need to stick a low value
> resistor
> > in the base. Done properly, they are very reproducible and reasonably
> > insensitive to load.
>
> Is that required for real circuits or just for the simulations?
>
>
> --
> These are my opinions.  I hate spam.
>
>
>
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Re: [time-nuts] Switching transistors, current sources, nonidealties and noise

[Bob Camp]

Hi

Works for both.

Bob

> On Jul 1, 2016, at 2:54 PM, Hal Murray  wrote:
> 
> 
> kb...@n1k.org said:
>> There is also the somewhat non-intuitive need to stick a low value resistor
>> in the base. Done properly, they are very reproducible and reasonably
>> insensitive to load.  
> 
> Is that required for real circuits or just for the simulations?
> 
> 
> -- 
> These are my opinions.  I hate spam.
> 
> 
> 
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