While the Brooks Shera (http://www.rt66.com/~shera/index_fs.htm) GPS OCXO Controller design is excellent for use with an HP 10811, there are challenges to using the standard design with low sensitivity oscillators. The Shera controller uses a 24MHz phase counter design with the gain set for a 7.5e-9 / volt sensitivity over a +/- 3 volt span, or a 4.5e-8 controller span. To interface the controller to a typical HP 10811 the output is fed through a divider to the bottom of a frequency trim pot. A precision reference voltage is fed to the top of the pot, and the oscillator EFC input is driven from the wiper. The frequency trim pot provides a fixed offset voltage to set the frequency, and this voltage is "modulated" by the controller output to maintain GPS lock. This design works well with oscillators capable of frequency spans of 1e-7, and sensitivities of > 1.5e-8/volt. Less sensitive oscillators require different methods to match the oscillator sensitivity to the controller span. In interfacing an Isotemp mil spec version of the HP 10811B with a sensitivity of 9.7e-9/volt and a span of 4.85e-8 to the Shera controller even with RA and RB removed and the DAC feeding the frequency trim pot directly the pot acts as a divide by 2 to the controller output; and additional gain would have been required to match the controller span. Adding an additional gain stage after the DAC output results in amplifying the DAC noise, which can adversely affect the short-term stability of the oscillator. By feeding an inverting summing amplifier and inverter with the offset and controller outputs and driving the oscillator from the inverter output, an effective gain of 2 was realized by removing the frequency trim pot from the controller path without adding additional gain to amplify the DAC noise. Using this arrangement allowed the lower sensitivity Isotemp oscillator to be matched to the 4.5e-8 controller span without adding additional gain. Rubidium oscillators can be used with the standard Shera design, but the low sensitivity (1e-9/volt for my LPRO, 4.5e-10/volt for my FRS-C) requires the gain to be 9x - 20x normal to match the controller span. (Up to 40x for the FRS-C with the original frequency pot arrangement!) Multiplying the DAC noise by a factor of 9x - 20x is not a good approach for maximizing short-term stability in a precision oscillator. The available span for the LPRO is 5e-9, so it can only use 12% of the available 4.5e-8 controller span if matched with an external gain stage. The FRS-C has a span of 2.25e-9, and can use only 5% of the controller span. Brooks will share his source code if you ask him nicely, so modifications to the controller software are possible. But just increasing the gain 2x in the controller software requires changing the limiter and restricting the span in the lower filter modes, and higher gains result in other design issues making an additional gain of 2x the practical limit, and only with additional work on the limiter. There are other options to achieve the required gain, and for rubidium oscillator control they are appropriate. One method to increase the gain is to use a faster clock and a shorter sample time. Operation of a similar design using 74F163 counters at speeds up to 125MHz is possible. Due to path delays encountered in high-speed operation, layout becomes a major factor for reliable operation at speeds above 50MHz. Standard perf-board techniques work well up to 50MHz, so a design was tested using two 74F163 counters, a 74HC165 shift register, and a 50MHz oscillator in the phase detector. The counter is read and reset each second and the results accumulated in the PIC. The result is scaled for the sample time chosen (divide by 2 for 60 sec) and fed to the process. The sample divider used was an ST 74HC393 dual binary counter, with one divider used to divide the 10MHz oscillator down to 625KHz to feed the phase detector and the second divider used to divide the 50MHz clock by 4 to supply a 12.5MHz clock for the PIC. The faster clock reduces the controller span from 4.5e-8 to 2.16e-8, still well beyond the span of a rubidium oscillator. It also increases the controller sensitivity to 3.6e-9/volt giving an effective gain of 2.08 over the original 24 MHz design. Another approach to increasing the effective gain is to increase the filter constant. This requires more stability in the oscillator to be effective, which is just what a rubidium oscillator offers. Each doubling of the filter constant effectively increases the gain by 2. Increasing the filter constant becomes the best way to get the gain required for a rubidium oscillator controller without adding an additional gain stage. Increasing the sample time to 60 sec results in a controller sensitivity of 9e-10/volt with a 5.4e-9 span using a 50 MHz clock. This configuration will properly drive an LPRO with 1e-9 / volt sensitivity using 93% of the available controller span. For the lower sensitivity FRS-C, the F1 filter constant is adjusted 1 step to scale the filter by an additional factor of 2. This results in a controller sensitivity of 4.5e-10 / volt and a span of 2.7e-9. This provides a proper match to the FRS-C sensitivity, using 83% of the available controller span. Because a rubidium oscillator has poorer short-term stability when compared to a good OCXO, short-term variations in EFC should be suppressed when using a rubidium oscillator, and the Shera design has an Alpha filter available to do just that. Once stable in your selected mode, the Alpha filter should be employed to maximize short-term stability. Keep in mind that the filter constant is now 2x or 4x longer than the stock controller so you are effectively starting the IIR filter in mode 3 or 4 and going up from there to effectively mode 8 or 9. The filter settle time for the highest modes become 1 to 2 days, and this can be too long to correct for daily variations in the environment. Settle times of about 12 hours produced the best long-term stability, with typical 1 day stability of < 5e-13 being achieved using a combination of these techniques on an LPRO.
----- Original Message ----- From: "Christopher Hoover" <[EMAIL PROTECTED]> To: <[email protected]> Sent: Sunday, September 18, 2005 10:40 AM Subject: [time-nuts] RE: phase locking Rb to GPS (was time-nuts Digest, Vol 14, Issue 30) > > On 9/18 "Tom Clark, W3IWI" <[EMAIL PROTECTED]> wrote: > > The frequency knob that you tweak to correct the Rb frequency > passes some tens of ma thru a coil surrounding the RF interaction > region. If you try to phase lock a Rb to GPS, you need to develop > a current source error signal. > > Hmmm ... can you say more about this current source error signal? > > I've been thinking of locking my Rb standard to GPS. I was simply going to > use the same circuit (1 pps, 10 MHz in; analog EFC voltage out) that I have > running with my OXCO but with different filter and EFC DAC coefficients. > > So is this not sufficient to phase lock GPS to a Rb standard? > > Naively, > -- Christopher. > > > _______________________________________________ > time-nuts mailing list > [email protected] > https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts _______________________________________________ time-nuts mailing list [email protected] https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
