[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

Ok, single mixer phase noise basics:

First thing is to womp the mixer up to the point it almost smokes. Putting +7 
dbm into
both ports on a “7 dbm” mixer is very normal in this case. Watching for the 
fact that the
mixer likely is *not* a 50 ohm load is part of the process ( = pads might help 
out) as well 
as understanding that it does not have a monster amount of isolation ( = 
isolation amps 
may be needed ).

Next one generates a beat note by offsetting the two oscillators a bit. This 
gives you a nice 
360 degree sweep function ( 360 degrees per cycle :) ). From that you can work 
out the 
system sensitivity in volts per degree ( or better yet per radian since that’s 
what you actually
want as the “magic number” …. it’s phase modulation so radian is king …).

Next you lock the two oscillators together via a DC feed out of the mixer to 
one or the 
other of them. You adjust the “lock point” so that it is at zero volts out of 
the mixer. This
puts the two oscillators in quadrature. Yes, there is that messy 2X the input 
frequency RF
output and the inevitable leakage. Those are handled with a lowpass filter. 

The output of the mixer is now “just noise”. There is no nasty carrier to deal 
with. There is 
no messy fold over to wonder about. What you get is the DSB noise ( so both 
sides of
carrier) from the sum of the two oscillators. 

Output of the mixer goes up if you terminate it in an “high” load. Something 
like 500 ohms
on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to 
be ok
either way and the added gain / better floor is “free”. 

Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 
3 db if the 
oscillators are identical. Your “output” is 6 db higher than the single 
sideband / single oscillator 
phase noise. You can argue that close in noise is likely correlated due to it 
being a modulation
on the carrier. The standard convention is to use 3 db.

Amplify the noise up and you can measure very low levels of phase noise. Low 
noise
audio op-amps are pretty easy to find spec sheets on. With anything these days 
finding
them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance 
in the 
feedback path go way back for this application. There are a lot of other 
candidates. 

The cutoff of the lock signal typically is adjustable to keep it below the 
lowest point 
of interest for your noise testing. If that is impractical, there are ways to 
calibrate and 
read “inside the loop”. 

The HP 3048 phase noise analyzer was based on this approach. The original app 
note most
folks started from came from Fluke back in the early 1970’s. I have not (yet) 
found a good
copy of it on the internet. 

Fun !!!

Bob



> On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Bob, Magnus,
> Thanks, clear. A counter is for ADEV, not for phase noise.
> I made a test setup to learn how to use the mixer/PLL approach.
> First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC 
> shift with shifting the phase between the two signal.
> Then by modulating one output with FM or PM.
> There is a low pass filter after the mixer to get rid of the 10 MHz and its 
> harmonics but the LPF is measured flat till about 10kHz.
> The output signal from the mixer was kept within 10% of the full voltage 
> swing to stay in the (hopefully) linear range.
> Using PM creates a low frequency output from the mixer that is proportional 
> to the phase shift (region 0-1 degree) and constant in amplitude with change 
> of frequency. Also when using external modulation from an audio signal 
> generator created the expected behavior with drive level and no frequency 
> impact
> Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases 
> very fast with increasing frequency (range 0.1 to 10 Hz)
> Also when using 1 Hz or more frequency deviation. The higher frequency 
> deviation leads to higher output levels as expected.
> Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation 
> and 0.1 to 1 Hz frequency deviation) translates to the calibration example 
> mentioned in the document on phase noise measurement as linked by Bob. (0.1 
> Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
> At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a 
> lot of output. Why is this output frequency dependency?
> Is this a problem with the signal generator?  Or the mixer?
> Then I tried to use the modulated signal from the SG PLL locked to a 10MHz 
> VCO. Results where the same. FM output signal is frequency dependent, PM not.
> Erik.
> 
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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Magnus Danielson via time-nuts]

Hi Jim,

On 2022-06-20 17:57, Lux, Jim via time-nuts wrote:

On 6/20/22 2:39 AM, Magnus Danielson via time-nuts wrote:



So, a counter is really like an ADC for phase, with wide bandwidth 
input and a sub-sampling mechanism (trigger/time-base). Through 
processing frequency estimates can be provided. Aliasing occurrs in 
the sub-sampling. Modern counters can provided estimation filters 
than goes from a higher sub-sampling rate to a lower, which to some 
degree removes aliasing, but not fully. These frequency estimation 
methods form a form of decimation filter.


Cheers,
Magnus 


An intruiging thought as I drink my first cup of coffee (meaning it's 
not well thought out)..

Enjoy!


jumping off from "counter is similar to an ADC for phase" - is there a 
time domain equivalent for Nyquist criterion?   Certainly there's the 
cycle ambiguity.. you know when the zerocrossing occurred, but not how 
many are in between (although a counter usually does). For everything 
else there is a frequency/time duality, so I suspect there is.  The 
criterion is usually explained in terms of information - so there 
should be an equivalent "has all the information" statement for 
counters/gate widths/precisions.


Well, considering that optimum phase/time sensitivity is at the 
through-zero of a sine, with the optimum slew-rate of the signal, you 
have two observation points per cycle. You can view that as having 
essentially two sample-points of phase per cycle. Similarly you will 
have two optimal sample-points for amplitude in quadrature on the peaks 
of the sine.


Now, using this fact, you have a Nyquistian type of relationship and 
also upper phase-information frequency being that of the cosine itself, 
since you can fit a modulaiton that pushes the rising edge one way and 
the falling edge the other way. As you attempt a higher modulation 
frequency you cannot distinguish that from the mirror frequency lower 
than that frequency. Thus, they Nyquist frequency of modulation is the 
carrier frequency.


But then again, the same can be said for any overtones, so you can 
support higher modulation frequencies there, with the same basic rule. 
However, sorting that out can be a bit tricky, considering non-linear 
functions and intermodulations.


PS. IEEE Std. 1139-2022 made it through a formal approval after 
balloting, so now it is off for last editorial touch-ups before 
publishing. Good news. Look forward to put it into use.


Cheers,
Magnus



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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Erik Kaashoek via time-nuts]

I found the info on how to calibrate and lots of other practical stuff in
here: https://martein.home.xs4all.nl/pa3ake/hmode/dds_pmnoise_pll.html
Erik


On Mon, Jun 20, 2022, 19:43 Erik Kaashoek  wrote:

> Bob, Magnus,
> Thanks, clear. A counter is for ADEV, not for phase noise.
> I made a test setup to learn how to use the mixer/PLL approach.
> First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the
> DC shift with shifting the phase between the two signal.
> Then by modulating one output with FM or PM.
> There is a low pass filter after the mixer to get rid of the 10 MHz and
> its harmonics but the LPF is measured flat till about 10kHz.
> The output signal from the mixer was kept within 10% of the full voltage
> swing to stay in the (hopefully) linear range.
> Using PM creates a low frequency output from the mixer that is
> proportional to the phase shift (region 0-1 degree) and constant in
> amplitude with change of frequency. Also when using external modulation
> from an audio signal generator created the expected behavior with drive
> level and no frequency impact
> Using FM with 0.1 Hz frequency deviation the mixer output amplitude
> decreases very fast with increasing frequency (range 0.1 to 10 Hz)
> Also when using 1 Hz or more frequency deviation. The higher frequency
> deviation leads to higher output levels as expected.
> Can someone help me understand how this FM signal (0.1 to 1000 Hz
> modulation and 0.1 to 1 Hz frequency deviation) translates to the
> calibration example mentioned in the document on phase noise measurement as
> linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband (at
> 1kHz?) level of -86 dBc)
> At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is
> a lot of output. Why is this output frequency dependency?
> Is this a problem with the signal generator?  Or the mixer?
> Then I tried to use the modulated signal from the SG PLL locked to a 10MHz
> VCO. Results where the same. FM output signal is frequency dependent, PM
> not.
> Erik.
>
>
>
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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Erik Kaashoek via time-nuts]

Bob, Magnus,
Thanks, clear. A counter is for ADEV, not for phase noise.
I made a test setup to learn how to use the mixer/PLL approach.
First using 10MHz from both outputs of a DSS (Rigol DG990) to observe 
the DC shift with shifting the phase between the two signal.

Then by modulating one output with FM or PM.
There is a low pass filter after the mixer to get rid of the 10 MHz and 
its harmonics but the LPF is measured flat till about 10kHz.
The output signal from the mixer was kept within 10% of the full voltage 
swing to stay in the (hopefully) linear range.
Using PM creates a low frequency output from the mixer that is 
proportional to the phase shift (region 0-1 degree) and constant in 
amplitude with change of frequency. Also when using external modulation 
from an audio signal generator created the expected behavior with drive 
level and no frequency impact
Using FM with 0.1 Hz frequency deviation the mixer output amplitude 
decreases very fast with increasing frequency (range 0.1 to 10 Hz)
Also when using 1 Hz or more frequency deviation. The higher frequency 
deviation leads to higher output levels as expected.
Can someone help me understand how this FM signal (0.1 to 1000 Hz 
modulation and 0.1 to 1 Hz frequency deviation) translates to the 
calibration example mentioned in the document on phase noise measurement 
as linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband 
(at 1kHz?) level of -86 dBc)
At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there 
is a lot of output. Why is this output frequency dependency?

Is this a problem with the signal generator?  Or the mixer?
Then I tried to use the modulated signal from the SG PLL locked to a 
10MHz VCO. Results where the same. FM output signal is frequency 
dependent, PM not.

Erik.

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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

A “proper” phase noise analyzer will get you down to < -165 dbc/Hz. It also 
will preserve
the frequency spectrum ( no sampling / nyquist roll over ). What you get at 132 
Hz offset 
is the (DSB) noise at that offset and *only* that noise. 

With a counter ( as Magnus mentions ) the sampling process looks at and sums up 
all the
noise in the input bandwidth. If it’s a 0.01 second sample, you get noise from 
100Hz, 200Hz,
300Hz on out to whatever the counter’s input bandwidth is. 

Since the counter likely has a bandwidth of a couple hundred MHz and you are 
measuring
something like 10 or 25 MHz, you sum up a whole lot of “noise floor” noise. 
Even if the counter
input amp has a zero db noise figure and the sampler does as well, you “fold 
in” 200 MHz worth
of that noise. This makes the sensitivity of the counter to low levels of phase 
noise pretty poor. 

If you didn’t have this sort of limitation, your counter might well have 20 fs 
resolution rather than 
20 ps …. (and no, this isn’t the only reason you have limited resolution)

Bob

> On Jun 19, 2022, at 10:45 PM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Bob,
> Many thanks for the guidance you provide and the phase noise measurement 
> document.
> Can you provide feedback on this reasoning: A counter is like an ADC but in 
> the frequency domain. So if you measure with 0.01 s tau you basically average 
> over 0.01 s so you can only observe "phase noise" (e.g. energy that is not at 
> the exact requested frequency) up to maximum 50 Hz from the carrier. But as 
> you measure the true frequency changes the sensitivity of this measurement is 
> extremely high. Translating the amount of time spend at a certain frequency 
> away from the carrier (ADEV?) into a phase noise number in dBc is something I 
> do not yet understand.
> With a (very good) spectrum analyzer you may be able to come close to the 
> carrier but as there is so much energy in the carrier it will be difficult to 
> observe phase noise energy closer than say 1 or 10 kHz (at least not with the 
> equipment I can afford) so any phase noise plot created using a spectrum 
> analyzer can not be better than the combined phase noise of all LO's in the 
> spectrum analyzer and will start at say 1 or 10 kHz.
> For the frequencies between 50 Hz and 20 kHz the simplest option is to use a 
> second LO and a mixer and a slow  (loop BW below 10 Hz)PLL to keep the mixer 
> in quadrature and feed the output of the mixer, after low pass filtering, 
> into a PC soundcard for FFT processing.
> Erik.
> 
> On 19-6-2022 22:45, Bob kb8tq via time-nuts wrote:
>> Hi
>> 
>> As HP found out back around 1973 or so, translating ADEV to phase noise
>> is not possible. This is true, even if you have the ADEV numbers for a 
>> variety
>> of Tau values as opposed to some sort of “average” kind of number.
>> 
>> There are a number of things ( like spurs ) that can strongly influence a 
>> counter
>> based ADEV reading, and have very little impact on a phase noise ( or signal 
>> to
>> noise reading.  There also are ways the shape of the phase noise curve can
>> impact ADEV and have very little signal to noise impact for a specific 
>> signal.
>> 
>> By far the best way to do this is to properly measure phase noise at various
>> offsets from carrier. You can then look at the dbc/Hz numbers at each offset.
>> This lets you see what your devices are doing to the signal. You can then 
>> track
>> down the offending bit or piece and fix the problem.
>> 
>> The easiest way I know of to do phase noise is to quadrature lock two 
>> identical
>> sources into a double balanced mixer. You then put in a simple amplifier 
>> stage
>> to drive the mix down output into a sound card or spectrum analyzer. Total 
>> cost
>> if you already have a sound card should be < $50 ( US dollars …) for a DIY 
>> version.
>> That assumes you have the usual junk box parts and do a point to point wire
>> version.
>> 
>> Some example ADEV plots:
>> 
>> http://leapsecond.com/museum/manyadev.gif 
>> 
>> 
>> http://leapsecond.com/museum/manyadev.gif 
>> 
>> 
>> Some plots of a number of measurements:
>> 
>> http://www.leapsecond.com/pages/fe405/ 
>> 
>> 
>> Quick primer on phase noise measurement
>> 
>> https://www.npl.co.uk/special-pages/guides/gpg68_noise 
>> 
>> 
>> ( The easy approach starts on page 21 :) )
>> 
>> Bob
>> 
>> 
>>> On Jun 19, 2022, at 11:40 AM, Karen Tadevosyan via time-nuts 
>>>  wrote:
>>> 
>>> Hi
>>> 
>>> 
>>> 
>>> Thank you for the clarification and rf-tools link.
>>> 
>>> 
>>> 
>>> Agree with your calculation. That’s why I raised this question regarding a 
>>> fixing PN degradation by Pendulum CNT-91.
>>> 
>>> 
>>> 
>>> Could you please explain where is the error in my reasoning of the 
>>> experiment :
>>> 
>>> 
>>> 
>>> *   There is one 10 MHz OCXO 

[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Lux, Jim via time-nuts]

On 6/20/22 2:39 AM, Magnus Danielson via time-nuts wrote:



So, a counter is really like an ADC for phase, with wide bandwidth 
input and a sub-sampling mechanism (trigger/time-base). Through 
processing frequency estimates can be provided. Aliasing occurrs in 
the sub-sampling. Modern counters can provided estimation filters than 
goes from a higher sub-sampling rate to a lower, which to some degree 
removes aliasing, but not fully. These frequency estimation methods 
form a form of decimation filter.


Cheers,
Magnus 


An intruiging thought as I drink my first cup of coffee (meaning it's 
not well thought out)..


jumping off from "counter is similar to an ADC for phase" - is there a 
time domain equivalent for Nyquist criterion?   Certainly there's the 
cycle ambiguity.. you know when the zerocrossing occurred, but not how 
many are in between (although a counter usually does). For everything 
else there is a frequency/time duality, so I suspect there is.  The 
criterion is usually explained in terms of information - so there should 
be an equivalent "has all the information" statement for counters/gate 
widths/precisions.


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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Magnus Danielson via time-nuts]

Erik,

A counter actually measures a number of phase measurements. Then, as you 
process that you get a frequency readout based on the difference between 
them (event-count divided by time between phase measurements). Now, as 
you want to do frequency read-out, you can do a handful of filtering 
mechanisms, and the CNT-91 can do the linear regression. This filtering 
takes a number of samples and provides a filter to estimate frequency. 
The consequence of that is that variations you see now have a different 
scale than if you did the original calculation of only two 
phase-samples. This creates a bias function and variations needs to be 
corrected for to get numbers you can relate to the normal scale. It's 
great for giving better frequency readings, but if you aim to quantify 
the variations you end up fooling yourself.


Also, your assumption on observation frequency in Nyquist is wrong. 
Turns out that aliasing of higher frequencies is very problematic. It is 
only very recent instruments that can have the ability to avoid aliasing 
(by using digital decimation), but a counter is not one of them, it is 
fully exposed to the aliasing problem.


There is translations charts to convert the noise-amplitude of each 
noise type into phase-noise and ADEV readings. If you have truely random 
noise obeying the rules, you can convert between them. Toss in a spur, 
and it works differently, and well, you need to convert those too 
according to other rules. Look for "Enricos chart".


Noise types reaching for high frequencies compared to measurement tau0 
will affect the resulting ADEV for sure. The bandwidth of that even 
affects white phase modulation directly, and flicker phase modulation to 
some degree.


So, a counter is really like an ADC for phase, with wide bandwidth input 
and a sub-sampling mechanism (trigger/time-base). Through processing 
frequency estimates can be provided. Aliasing occurrs in the 
sub-sampling. Modern counters can provided estimation filters than goes 
from a higher sub-sampling rate to a lower, which to some degree removes 
aliasing, but not fully. These frequency estimation methods form a form 
of decimation filter.


Cheers,
Magnus

On 2022-06-20 08:45, Erik Kaashoek via time-nuts wrote:

Bob,
Many thanks for the guidance you provide and the phase noise 
measurement document.
Can you provide feedback on this reasoning: A counter is like an ADC 
but in the frequency domain. So if you measure with 0.01 s tau you 
basically average over 0.01 s so you can only observe "phase noise" 
(e.g. energy that is not at the exact requested frequency) up to 
maximum 50 Hz from the carrier. But as you measure the true frequency 
changes the sensitivity of this measurement is extremely high. 
Translating the amount of time spend at a certain frequency away from 
the carrier (ADEV?) into a phase noise number in dBc is something I do 
not yet understand.
With a (very good) spectrum analyzer you may be able to come close to 
the carrier but as there is so much energy in the carrier it will be 
difficult to observe phase noise energy closer than say 1 or 10 kHz 
(at least not with the equipment I can afford) so any phase noise plot 
created using a spectrum analyzer can not be better than the combined 
phase noise of all LO's in the spectrum analyzer and will start at say 
1 or 10 kHz.
For the frequencies between 50 Hz and 20 kHz the simplest option is to 
use a second LO and a mixer and a slow  (loop BW below 10 Hz)PLL to 
keep the mixer in quadrature and feed the output of the mixer, after 
low pass filtering, into a PC soundcard for FFT processing.

Erik.

On 19-6-2022 22:45, Bob kb8tq via time-nuts wrote:

Hi

As HP found out back around 1973 or so, translating ADEV to phase noise
is not possible. This is true, even if you have the ADEV numbers for 
a variety

of Tau values as opposed to some sort of “average” kind of number.

There are a number of things ( like spurs ) that can strongly 
influence a counter
based ADEV reading, and have very little impact on a phase noise ( or 
signal to
noise reading.  There also are ways the shape of the phase noise 
curve can
impact ADEV and have very little signal to noise impact for a 
specific signal.


By far the best way to do this is to properly measure phase noise at 
various
offsets from carrier. You can then look at the dbc/Hz numbers at each 
offset.
This lets you see what your devices are doing to the signal. You can 
then track

down the offending bit or piece and fix the problem.

The easiest way I know of to do phase noise is to quadrature lock two 
identical
sources into a double balanced mixer. You then put in a simple 
amplifier stage
to drive the mix down output into a sound card or spectrum analyzer. 
Total cost
if you already have a sound card should be < $50 ( US dollars …) for 
a DIY version.
That assumes you have the usual junk box parts and do a point to 
point wire

version.

Some example ADEV plots:

[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Karen Tadevosyan via time-nuts]

Hi

 

Bob, many thanks for your explanations/recommendations and links.

 

According your advices I will try to make PN measurement on new SA with 
cross-correlation to get more clear picture of the output signals.

 

The next my step will be a quadrature method measurement  and comparison of the 
results.

 

Karen, ra3apw 

 

From: Bob kb8tq  
Sent: Sunday, June 19, 2022 11:46 PM
To: Discussion of precise time and frequency measurement 

Cc: Karen Tadevosyan 
Subject: Re: [time-nuts] Fixing PN degradation via ADEV measurement

 

Hi

 

As HP found out back around 1973 or so, translating ADEV to phase noise 

is not possible. This is true, even if you have the ADEV numbers for a variety

of Tau values as opposed to some sort of “average” kind of number.

 

There are a number of things ( like spurs ) that can strongly influence a 
counter

based ADEV reading, and have very little impact on a phase noise ( or signal to

noise reading.  There also are ways the shape of the phase noise curve can

impact ADEV and have very little signal to noise impact for a specific signal. 

 

By far the best way to do this is to properly measure phase noise at various 

offsets from carrier. You can then look at the dbc/Hz numbers at each offset. 

This lets you see what your devices are doing to the signal. You can then track

down the offending bit or piece and fix the problem. 

 

The easiest way I know of to do phase noise is to quadrature lock two identical

sources into a double balanced mixer. You then put in a simple amplifier stage

to drive the mix down output into a sound card or spectrum analyzer. Total cost

if you already have a sound card should be < $50 ( US dollars …) for a DIY 
version.

That assumes you have the usual junk box parts and do a point to point wire

version. 

 

Some example ADEV plots:

 

http://leapsecond.com/museum/manyadev.gif

 

http://leapsecond.com/museum/manyadev.gif

 

Some plots of a number of measurements:

 

http://www.leapsecond.com/pages/fe405/

 

Quick primer on phase noise measurement 

 

https://www.npl.co.uk/special-pages/guides/gpg68_noise

 

( The easy approach starts on page 21 :) )

 

Bob

 

 

On Jun 19, 2022, at 11:40 AM, Karen Tadevosyan via time-nuts 
mailto:time-nuts@lists.febo.com> > wrote:

 

Hi 



Thank you for the clarification and rf-tools link.



Agree with your calculation. That’s why I raised this question regarding a 
fixing PN degradation by Pendulum CNT-91.



Could you please explain where is the error in my reasoning of the experiment :



*  There is one 10 MHz OCXO with ADEV = 5 mHz
*  There are two boards (DUT1 and DUT2) which multiply 10 MHz OCXO 
signal by 2.5 using the PLL method
*  DUT1 has 25 MHz output signal with high PN  (checking by air and by 
measurement of S/N)
*  DUT2 has 25 MHz  output signal with low PN  (checking by air and by 
measurement of S/N)
Experiment’s steps:
*  Step 1: DUT1 ADEV measuring gives me a value of 60 - 70 mHz instead 
of the expected 12.5 mHz  (5 mHz x 2.5)
*  Step 2: DUT2 ADEV measuring gives me a value of 10 - 12 mHz which 
matches the expected 12.5 mHz  (5 mHz x 2.5)
*  Step 3: based on ADEV values which in the first case (DUT1) are much 
greater than expected and in the second case (DUT2) coincide with the expected 
I conclude that PN of the output signal from DUT2 will be lower than from DUT1.
I can see this PN degradation using Pendulum CNT-91 only as R FSQ8 does not 
fixate any PN degradation between DUT1 and DUT2

Karen, ra3apw

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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Erik Kaashoek via time-nuts]

Bob,
Many thanks for the guidance you provide and the phase noise measurement 
document.
Can you provide feedback on this reasoning: A counter is like an ADC but 
in the frequency domain. So if you measure with 0.01 s tau you basically 
average over 0.01 s so you can only observe "phase noise" (e.g. energy 
that is not at the exact requested frequency) up to maximum 50 Hz from 
the carrier. But as you measure the true frequency changes the 
sensitivity of this measurement is extremely high. Translating the 
amount of time spend at a certain frequency away from the carrier 
(ADEV?) into a phase noise number in dBc is something I do not yet 
understand.
With a (very good) spectrum analyzer you may be able to come close to 
the carrier but as there is so much energy in the carrier it will be 
difficult to observe phase noise energy closer than say 1 or 10 kHz (at 
least not with the equipment I can afford) so any phase noise plot 
created using a spectrum analyzer can not be better than the combined 
phase noise of all LO's in the spectrum analyzer and will start at say 1 
or 10 kHz.
For the frequencies between 50 Hz and 20 kHz the simplest option is to 
use a second LO and a mixer and a slow  (loop BW below 10 Hz)PLL to keep 
the mixer in quadrature and feed the output of the mixer, after low pass 
filtering, into a PC soundcard for FFT processing.

Erik.

On 19-6-2022 22:45, Bob kb8tq via time-nuts wrote:

Hi

As HP found out back around 1973 or so, translating ADEV to phase noise
is not possible. This is true, even if you have the ADEV numbers for a variety
of Tau values as opposed to some sort of “average” kind of number.

There are a number of things ( like spurs ) that can strongly influence a 
counter
based ADEV reading, and have very little impact on a phase noise ( or signal to
noise reading.  There also are ways the shape of the phase noise curve can
impact ADEV and have very little signal to noise impact for a specific signal.

By far the best way to do this is to properly measure phase noise at various
offsets from carrier. You can then look at the dbc/Hz numbers at each offset.
This lets you see what your devices are doing to the signal. You can then track
down the offending bit or piece and fix the problem.

The easiest way I know of to do phase noise is to quadrature lock two identical
sources into a double balanced mixer. You then put in a simple amplifier stage
to drive the mix down output into a sound card or spectrum analyzer. Total cost
if you already have a sound card should be < $50 ( US dollars …) for a DIY 
version.
That assumes you have the usual junk box parts and do a point to point wire
version.

Some example ADEV plots:

http://leapsecond.com/museum/manyadev.gif 


http://leapsecond.com/museum/manyadev.gif 


Some plots of a number of measurements:

http://www.leapsecond.com/pages/fe405/ 

Quick primer on phase noise measurement

https://www.npl.co.uk/special-pages/guides/gpg68_noise 


( The easy approach starts on page 21 :) )

Bob



On Jun 19, 2022, at 11:40 AM, Karen Tadevosyan via time-nuts 
 wrote:

Hi



Thank you for the clarification and rf-tools link.



Agree with your calculation. That’s why I raised this question regarding a 
fixing PN degradation by Pendulum CNT-91.



Could you please explain where is the error in my reasoning of the experiment :



*   There is one 10 MHz OCXO with ADEV = 5 mHz
*   There are two boards (DUT1 and DUT2) which multiply 10 MHz OCXO signal 
by 2.5 using the PLL method
*   DUT1 has 25 MHz output signal with high PN  (checking by air and by 
measurement of S/N)
*   DUT2 has 25 MHz  output signal with low PN  (checking by air and by 
measurement of S/N)
Experiment’s steps:
*   Step 1: DUT1 ADEV measuring gives me a value of 60 - 70 mHz instead of 
the expected 12.5 mHz  (5 mHz x 2.5)
*   Step 2: DUT2 ADEV measuring gives me a value of 10 - 12 mHz which 
matches the expected 12.5 mHz  (5 mHz x 2.5)
*   Step 3: based on ADEV values which in the first case (DUT1) are much 
greater than expected and in the second case (DUT2) coincide with the expected 
I conclude that PN of the output signal from DUT2 will be lower than from DUT1.
I can see this PN degradation using Pendulum CNT-91 only as R FSQ8 does not 
fixate any PN degradation between DUT1 and DUT2

Karen, ra3apw

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