Bob Camp wrote:
Hi
Again more or less in order:
I'm trying to keep things as simple as I can at least to start. That rules out
the clean up loop oscillator at least in the beginning. It is a good idea, and
eventually I'll probably put one in.
I guess I'm going to need to do some looking on transformer feedback high
isolation amps. Everything I've seen so far on hight isolation has been
straight / no feedback stuff.
John Miles did some phase noise measurements
(http://www.thegleam.com/ke5fx/norton.htm) on designs like those at:
http://www.ko4bb.com/~bruce/CE_TransformerFeedback_BufferAmplifier.html
<http://www.ko4bb.com/%7Ebruce/CE_TransformerFeedback_BufferAmplifier.html>
John and I actually used 2N5109s and 2N5943's respectively.
I have LTSpice simulation files for some of these designs (however I
don't have a spice model for either transistor).
If you need more isolation just cascade a few such stages.
Simulation indicates that such stages can easily produce an output of
+23dBm or more should you need it.
The reverse isolation of a single stage is about 40dB which is easily
measured with a scope.
The collector current of such a CE stage is significantly lower than
that of a CB stage with similar output and distortion.
The input distortion of a CB stage limits the distortion performance
unless one augments the circuit with another transistor or a transformer.
Then collector output capacitance modulation limits the distortion
performance especially with high collector impedances.
The loading at RF on the mixer does reduce the audio output, but it improves
the isolation / match on the mixer. You trade one for the other.
The matching requirement is a red herring (there are various HP/Agilent
and Watkins-Johnson application notes on reducing mixer noise and loss
by reflecting all the unwanted mixer products back into the mixer). NIST
also did some work on the advantages of a capacitive mixer IF port
termination.
Resistors in series with the input will largely fix the matching and
locating the output stage of the isolation amplifier close to the mixer
also helps.
One way of reducing the mixer phase shift tempco (NIST claim a factor of
about 10) is to use it with the RF port unsaturated, however this
increases the noise.
The noise disadvantage can be offset by using a high level mixer.
Choose a mixer with high isolation as this usually indicates good diode
match and low transformer imbalance.
Usually DMTDs saturate both the IF and RF mixer ports.
Looks like some kind of local temperature stabilization might be a good idea
for the audio band limiting stuff. It's after the down convert, but some of the
time constants are indeed very long.
One can easily obtain 0.22uF NPO/C0G caps so one could parallel a few of
these and use low tempco resistors.
I suspect that silica dielectric cable is outside the budget constraints on
this project. Cheap foam coax in a spool on the floor or tiny stuff in the box,
possibly with better temperature control are about the only two choices.
Yes, NIST found it was outside their budget as well.
---------
Another very real choice is to simply move the goal post a bit. Pushing the
1x10*-12 point to 10 seconds from 1 second could turn out to be the only
economical basement alternative.
Off to bed ....
Bob
Bruce
On Jan 24, 2010, at 10:26 PM, Bruce Griffiths wrote:
Bob Camp wrote:
Hi
More or less in order:
The beat frequency is coming out of a rubidium. Hopefully it's fairly stable.
It won't be super quiet for 1 or .1 second tau. It looks like the counter will
be a FPGA time tagger, so the beat note frequency will drop out for free.
A cleanup loop may be useful to improve the offset source short term stability.
The cancellation of offset oscillator noise in a DMTD is imperfect.
The isolation amps are common base buffers. Not much gain, but quite a bit of
isolation. Phase shift / C - need to look into that.
You can achieve similar isolation together with lower noise and distortion with
a transformer feedback CE stage.
Transformer feedback CB stages have even lower noise coupled with low
isolation, however they can be useful for amplifying low level signals ahead of
a high isolation amplifier.
Mixer loading likely would be as I've done it before. Resistive termination at
RF and fairly high impedance at audio. Resistor here and there to improve the
match at RF. LC filtering adequate to suppress the RF stuff on the output of
the mixer. Single pole R-C for audio bandwidth control. Big capacitors and
small resistors for low noise.
That's one of the worst terminations possible from the noise perspective.
To lower the noise its essential to reflect the sum frequency back into the
mixer.
Resistors in series with the mixer LO and RF inputs will then be required to
improve the mixer input VSWR.
Until I've measured them I'm not sure of the floor of the limiters. Before I
get into that I want to be fairly sure I'm not over spec'ing them. If 100 ns is
as good as 3 ns it's not as hard a problem.
You can take the published phase noise for unspecified mixers as a lower limit.
The noise in the flicker region for the mixers (eg those from minicircuits)
that use integrated diode quads may be somewhat higher.
Initial measurements on a HP10534B (uses discrete diodes) appear consistent
with the typical noise specs for a low level mixer.
The issue of the group delay is an interesting one. I believe that people have
been getting good results with coax line for the phase shift. I'm a bit
conflicted on the coax. 15 meters of small diameter stuff will fit in the box
(maybe), but it's not super stable.. If I go foam coax then the phase shifter
gets pretty big. If I go with some kind of LC setup, temperature stability
would likely be an issue.
NIST's measurements indicate that lowest delay tempco is achieved with a
powdered silica dielectric.
Specialised fibres can have very low delay tempcos.
Crazy Stuff ....
So what did I miss that time?
Bob
Bruce
On Jan 24, 2010, at 9:01 PM, Magnus Danielson wrote:
Bob Camp wrote:
Hi
I realize that this is a bit off topic from the flow of the last few days. I
can only claim temporary insanity. Any comments about the temporary modifier in
that sentence being unneeded will of course be ignored...
Assuming that:
1) I have a DMTD setup of the "basement engineering" variety.
2) The beat note is> 5 Hz and< 10 Hz
3) The DUT's are all worse than 1x10^-12 at one second tau (no hydrogen masers
in the basement)
4) The offset oscillator is at least 2x10^-11 at one second tau.
5) The DUT's all put out 10 MHz
6) My counter will resolve 10 ns (= I could do better)
7) The limiters are good enough to not be an issue relative to the counter's 10
ns.
8) The zero crossings are phase shifted to be close, but not so close I arm
after I start during a run. 9) Regardless of the tau involved, nothing I'm
looking at will be better than 1x10-14
My down conversion from 10 MHz to 10 Hz gives me a 10^6 multiplication.
10 ns is a part in 10^8 at one second. It's a part in 10^7at 0.1 second (10 Hz).
First order, I should be able to hit (7+6 = 13) a part in 10^13 at less than 1 second.
That's significantly better than the DUT's. I don't need anything better in the counter
or limiters to measure what I'm looking at. Even if the limiters are 2X worse than the
counter, I'm still at the don't need better level in terms of counter and limiters. The
offset oscillator is going to cause some second order issues regardless of the limiters
and counter, but it still should be "ok". Next up:
If I phase shift one of the DUT's by 360 degrees, the beat note does the same. All I
need is 100 ns of phase shift to get everything lined up. I could do it with 180
degrees of shift and an phase inversion switch. I'm assuming (phase shifter and DMTD
stuff) can fit it all in a 2x4x8" box - I don't need a new bench to hold it
all ...
So what did I miss?
Remember that you *must* measure the actual beat frequency, since you will need
that to calculate the beat-gain. If it is between 5 and 10 Hz
the for a 10 MHz source your gain is 2E6 and 1E6 respectively, which is a
factor of 2 difference or 6 dB. So, your measurements will be inprecise from
that factor alone by +/- 3 dB. The remedy is fairly easy to come up with,
measure the input frequency and beat frequency for each arm. The best thing is
naturally to ensure that the beat frequencies of both arms is fairly close. EFC
steering of either source may work well enought in open-loop mode during
measurement (with the added benefit of not do spectral interference with the
phase noise which locked loop does).
How do you control the input levels to the mixers?
Do you have any isolational amplifiers?
How do you load and pre-filter the mixer outputs?
You haven't convinced me of the expected performance of the limiters.
I'm not sure it will be your biggest problem, but the way you phase-shift can
be of importance for the decorrelation loss.
Phase-shifting such that group-delay moves noise in time will be problematic,
since then the decorrelation gain of having phases coincide will be partly
lost since it is the group-delayed variant of the transfer oscillator against
the current-time transfer-oscillator (both delayed by each detector arm, but
only differnces is important). Vector-adding phase delays could work around
that. The optimum delay setting for cancelation may not be to fully
phase-adjust the leading edge.
That is what just popped up in my head at least.
Cheers,
Magnus
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