[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

HI

We’re not building a synthesizer here. We are putting together
a simple piece of test gear. The purpose of the test gear is to 
measure phase noise down into the -170 dbc / Hz range …..

Bob

> On Jul 12, 2022, at 4:05 PM, Mike Monett via time-nuts 
>  wrote:
> 
> To Bob kb8tq:
> 
> Unfortunately, most of your post made very little sense. D-flops are noisy,
> and the higher you go in frequency, the noisier they are. This is clear
> from the schematic. Here is the schematic for a MC1670 D-flop: MC1670SC.PDF.
> 
> Most of the noise is generated in the input SR latch. When the clock signal
> arrives, the input latch state is transferred to the output SR latch.
> Obviously, the sheer number of transistors involved is going to generate
> noise.
> 
> Unfortunately, the D-flop is needed in every known synthesizer. Keeping
> this noise out of the signal is the goal of every designer. Stanford
> Research Systems is one company that has mastered the art. See
> 
> https://www.thinksrs.com/products/siggen.html
> 
> But you need to know how much noise is involved. That is where my new
> method can help. I am busy collecting parts - the HMC984LP4E's will arrive
> tomorrow, and I am looking for a pair of low noise VCXO's.
> 
> I will have to regenerate test equipment that I haven't used in 5 decades
> to measure deadband, loop bandwidth, damping, crosstalk, jitter response,
> etc. I will also get a DBM to compare the results.
> 
> I will also need ripple filters for the electronics. I have described this
> before in 2N3906G.PNG
> 
> All this will take time - maybe months. But I have intended on making my
> own phase noise analyzer for a long time, and this will be an excellent way
> to get started.
> 
> Along the way, there are plenty of other projects to attend to: a 4GHz to
> 8GHz low noise signal generator using YIG oscillators, a GPSDO to supply an
> accurate 10MHz reference, a new method of eliminating the sampling jitter
> in the 1PPS signal from the GPDSO, a low noise VCXO to supply a 10MHz
> reference from the 1PPS signal, an ultra stable signal generator to allow
> sampling of signals up to 150GHz, and so on.
> 
> As there is little else to talk about, I will go silent while I am working
> on these projects.
> 
> Bye
> 
> Mike
> <2N3906G.PNG>___
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[time-nuts] Re: GPS failed

[Bob kb8tq via time-nuts]

Hi

The “typical” Symmetricom cone shaped GPS antennas are targeted at 
cell phone tower applications. Being mounted on the same structure
as multiple cell transmitters puts them in a significant RF environment. 
They have a *lot* of filtering built into the antenna to try to prevent overload
issues. 

As with a lot of things, Symmetricom simply rebrands antennas made by
others. Not all cone shaped antennas are identical. However it’s a pretty
good bet that most of them are very similar to what Symmetericom ( and
the other folks ) supply for cell applications. 

The “other end” of the range are the multi band saucer shaped “survey” 
antennas. They tend to have a lot less filtering and be more focused at 
allowing the user to access a wide range of frequencies ( both GNSS and
supplemental services) via a single device. Lots of filtering also tends to 
mess up delay here or there, that’s not a great thing for high precision 
work. 

Bob

> On Jul 12, 2022, at 9:39 AM, Mark Spencer via time-nuts 
>  wrote:
> 
> For what it is worth...
> 
> I have a commercial grade ( Symmetricom ?) GPS antenna on the roof of my 
> home.  I don't recall ever having any issues with GPS reception despite 
> having / had various other transmit / receive antennas on the roof for 
> various frequencies from 1.8 MHz thru 1.3 GHz.  Power levels on some bands 
> (not including 1.2 GHz thru 1.3 GHz where I have never exceeded approx 10 
> watts) can equal or occasionally exceed 100 watts.
> 
> As far as I know all my GPS receivers are using the typical 1.5 GHz GPS band.
> 
> As usual the experiences of others may differ from mine.
> 
> Best regards 
> Mark Spencer
> 
>> On Jul 12, 2022, at 12:08 AM, Matthias Welwarsky via time-nuts 
>>  wrote:
>> 
>> Hi,
>> 
>> if you're worried about in-band interference, the 23cm HAM radio band is 
>> reasonably close to the L1 GPS frequency. When I was still active in packet 
>> radio back in the days, our digipeater DB0DAR lost an interlink due to 
>> interference with a precision GPS receiver in use by another university 
>> institute. We had to shut it down. I think they operated a DGPS site at the 
>> time and our link traffic caused errors in the correction data. Or something.
>> 
>> BR,
>> Matthias
>> 
>>> On Montag, 11. Juli 2022 01:19:18 CEST skipp Isaham via time-nuts wrote:
>>> Hello to the Group,
>>> 
>>> I'd like to get some opinions and war stories regarding GPS reliability at
>>> high RF level and elevation locations.
>>> 
>>> Background:  Three different hill-top GPS receivers, all different types,
>>> using different antennas mounted on an outside fixiture, plain view of the
>>> open sky, all stopped working.
>>> 
>>> Test antennas were brought in and placed on a fixture well away from the
>>> original antennas, the recevers went back in to capture and lock.
>>> 
>>> From what I understand, the original antennas are what I would call straight
>>> preamp with no pre-selection / filtering.
>>> 
>>> The ordered and now inbound replacements are said to contain a SAW filter
>>> system. It is the intent of the client to just place these "improved
>>> antennas" in to service and get on with life.
>>> 
>>> I would suspect a GPS antenna (and receiver) could be subject to RF overload
>>> or blocking, however, we're assuming nothing major has changed at the site,
>>> nor any nearby location.  One might think there are more GPS receivers
>>> being pushed out of reliable operation by the world around them, I'm just
>>> not hearing those stories from a lot of people using them (GPS receivers).
>>> 
>>> Any new install GPS receiver antenna ordered will/should contain some
>>> pre-selection to potentially avoid a problem, even some years down the
>>> road? Seems like that's where things are going... no more off the shelf,
>>> wide band, (hot) preamplified GPS antennas in busy locations?
>>> 
>>> Thank you in advance for any related comments and/or opions ...
>>> 
>>> cheers,
>>> 
>>> skipp
>>> 
>>> skipp025 at jah who dot calm
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>> 
>> 
>> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik, Kaashoek)

[Bob kb8tq via time-nuts]

Hi

If you have any ceramic capacitors in the mix, they are often microphonic. 
The X7R versions are typically the best “high C” types. NPO’s normally are
completely non-microphonic. Other non-ceramic caps should be ok, but 
who knows. 

Roughly speaking, 1 nV / Hz should be low enough to not matter. Since all
these specs are “typical” one never knows quite what this or that part may
be doing. You *should* see a drop putting in a 1 nV in place of a 5 nV.

Bob

> On Jul 12, 2022, at 7:53 AM, Erik Kaashoek  wrote:
> 
> I'm struggling with the noise floor.
> First tests where done with a 5nV/sqrt(Hz) opamp. Noise floor with shorted 
> mixer output at 10kHz was -140dBc/Hz. Then I tried with 1nV/sqrt(Hz) opamp, 
> but that made no difference, noise floor at 10kHz was still -140dBc/Hz
> The setup was simplified to this schematic: 
> http://athome.kaashoek.com/time-nuts/PNA/SSPNA.JPG
> The REF_buffer creates a virtual ground, the Audio_LNA amplifies into the 
> differential audio output .
> Why did the lower noise opamp not make a difference?
> Also the setup is acting like a nice microphone. Tapping the housing is 
> clearly audible. Which component may be causing the microphony?
> 
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik, Kaashoek)

[Bob kb8tq via time-nuts]

Hi

There are a very small number of signal generators that *might* 
help when measuring phase noise on a good source. The “rest 
of them” are very much in the “don’t bother” category. Just which
one is in the “maybe” category depends a lot on your frequency
of interest. None of them seem to be very affordable …. wonder
why :) :) :)

Do useful ones pop up from time to time? Sure they do. They might
require a bit of work to get going. Thanks very much to those who
make them available. 

Bob

> On Jul 11, 2022, at 6:41 PM, Alex Pummer via time-nuts 
>  wrote:
> 
> yes there are much better signal generators out there, that frequency doubler 
> tuning circuit is for religious people only -- you need to be able to 
> believe, that it could work
> 73
> KJ6UHN
> Alex
> 
> On 7/11/2022 12:24 PM, Dave B via time-nuts wrote:
>> On 11/07/2022 08:30, time-nuts-requ...@lists.febo.com wrote:
>>> I also measured a Marconi 2022 signal generator and it was possible to
>>> lock but the phase noise was terrible with strong factional PLL spurs.
>> 
>> Indeed, those signal generators are renown for having "some rather 
>> interesting" spectral content...
>> Around, above and often well below the "desired" signal!
>> 
>> Not entirely surprising though, if you look at the block diagram of one such.
>> 
>> Regards to All.
>> 
>> Dave G0WBX(G8KBV)
>> 
>> 
>> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

Regardless of what you call the “ 1 Hz normalized noise “ of a digital
phase detector, it does predict what the noise floor does on it as the
reference frequency is changed over some reasonable range. This has
been demonstrated a lot of times and on a lot of different parts. 

Based on a number of RF designs using them ( and using gates for RF 
purposes ) the basic gate is what is at fault here. They are noisy and that 
noise changes with frequency. Frequency goes up / noise goes up. There
are very good reasons for this. 

Getting a gate with a noise figure below 6 db is highly unlikely …. That 
is what you would have to do in order to make a gate based circuit measure
a lower noise floor than the DBM based approach. Folks have spent a lot
of time searching for the magic “zero noise gate”. 

The sine wave component present at the DBM output at 2X the input 
frequency ( in the case of the phase noise test setup) are *way* higher
than the highest noise you are after. You put in 10 MHz or 100 MHz and
you go up to *maybe* 100 KHz on the noise. With a sound card, even 
getting to 100 KHz is going to be a challenge. 20 KHz may be the max. 

Knocking down the 2 x Fin component with a low pass filter is pretty easy.
Indeed the sound card or audio spectrum analyzer likely has some filtering 
already. The design and implementation of an adequate LPF is far from the 
biggest challenge that the person building the circuit will face. 

Indeed 1/F noise and noise corners do matter. All of the above has been
simply talking about noise floor. Gates have significant 1/F issues along
with their other “features”. This carries over to the detectors based on
them. As the gate speed goes up ( and the floor typically comes down),
the 1/F corner normally moves up ….

Bob

> On Jul 11, 2022, at 8:05 AM, Mike Monett via time-nuts 
>  wrote:
> 
> To Bob kb8tq
> 
>  Figure Of  Merit  sounds like a useless number. I  have  a different
>  approach that yields immediate and useful results. Before  I explain
>  my method, let me introduce myself.
> 
>  In  1970,   I   invented,   and   Memorex   patented,   the original
>  zero-deadband phase-frequency detector. You can see it in page  3 of
>  my '234 patent at https://patents.google.com/patent/US3810234A/
> 
>  This invention  soon   led   to   another   invention  of tremendous
>  significance to today's world.
> 
>  In 2014, researchers published a study in the journal Supercomputing
>  Frontiers and  Innovations  estimating the storage  capacity  of the
>  Internet at 1e24 bytes, or 1 million exabytes.
> 
>  When I  started  working for Memorex, an IBM  2314  disk  pack could
>  store 29.2  million  bytes.  At that  rate,  today's  internet would
>  require 1e24/29e6=3.44e16,  or 34,400,000,000,000,000 IBM  2314 disk
>  drives. This  is an impossible number. Other estimates  give equally
>  outrageous numbers.
> 
>  The problem  in those days was improvements in  disk  drive capacity
>  were basically  trial and error. This is a slow  and  very expensive
>  business.
> 
>  My new  invention allowed peering into the hard disk  and separating
>  out all   the   variables   that   affect   performance.   With this
>  information, researchers could see the effect of changes and quickly
>  optimize the performance. This allowed the tremendous improvement in
>  tape and  disk drive capacity that now allows the internet  to store
>  all the needed data.
> 
>  You can  see  how  this   invention   works  in  the  Katz  paper at
>  https://tinyurl.com/2bmuz3n2
> 
>  Now for my new method. 
> 
>  The schematic   for   a   phase-frequency   detector   is   shown in
>  DBAND2S.PNG. In  operation, a pulse arrives at the DATA pin  and pin
>  U1Q goes high. Then a pulse arrives at the VCO pin and pin  U2Q goes
>  high.
> 
>  This allows  the  NAND  gate  to bring  the  CLR  signal  low, which
>  immediately resets both d-flops.
> 
>  The result  is shown in ZERODB.PNG. It is a very  narrow  pulse with
>  both d-flops superimposed.
> 
>  This is the basis for my new approach. Simply tie both inputs of the
>  PFD together  and  measure  the noise spectrum  of  the  output. (Of
>  course, you have to ensure that both outputs match at zero error.)
> 
>  Once you have the PFD noise, you can enable the loop and measure the
>  total noise  spectrum. Then simply subtract the PFD spectrum  to get
>  the OCXO  noise.  If  you   have   two  identical  VCXO's,  each one
>  contributes half the noise.
> 
>  I don't know if this method would work with a double-balanced mixer.
>  The problem is a DBM requires quadrature signals, so the noise  is a
>  function of the OCXO noise as well as the mixer diodes. But the OCXO
>  noise is what you are trying to measure.
> 
>  This method  works with the PFD s

[time-nuts] Re: GPS failed

[Bob kb8tq via time-nuts]

Hi

These days there are folks who make a living tracking down interference
sources in the vicinity of ports and airports on a contract basis. Many of
the issues are navigation related. Some of it is GPS. Some is other stuff 
( like 5 GHz WiFI and radar …)

Bob

> On Jul 11, 2022, at 1:28 PM, Oz-in-DFW via time-nuts 
>  wrote:
> 
> On 7/11/2022 8:43 AM, Andy Talbot via time-nuts wrote:
>> I also heard a case of a GPS antenna going unstable, oscillating and taking
>> out most of the boats in a marina.
> There have also been several cases of cheap active TV antennas doing the same 
> thing. There was a case 10 or more years ago that shut down an LA marina and 
> the Port of Long Beach for several days until it was located.
> 
> Oz (in DFW, Texas near the airport)
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[time-nuts] Re: GPS failed

[Bob kb8tq via time-nuts]

Hi

Are all the receivers the same type / model? If so what are they? Various 
receivers made over the last 20 years have some “issues” that can pop up. 

Is the antenna gain properly matched to the needs of the receiver? Some
are designed for a “target gain” of 20 db, others 30, some 50. Match a 
20 db receiver with a 50 db antenna …. you have issues. 

Are all the hilltops in common view of each other? Somebody jamming their
ankle bracelet could be the simple answer if they are ….

Just what overloads this or that receiver often is a bit obscure. Most are 
pretty 
good at taking out CW tones. It often takes something a bit more complex
to drive them insane. This makes testing a bit exciting. 

Lots of strange and weird possibilities. 

Bob

> On Jul 10, 2022, at 3:19 PM, skipp Isaham via time-nuts 
>  wrote:
> 
> Hello to the Group, 
> 
> I'd like to get some opinions and war stories regarding GPS reliability at 
> high RF level and elevation locations. 
> 
> Background:  Three different hill-top GPS receivers, all different types, 
> using 
> different antennas mounted on an outside fixiture, plain view of the open 
> sky, 
> all stopped working. 
> 
> Test antennas were brought in and placed on a fixture well away from the 
> original antennas, the recevers went back in to capture and lock. 
> 
> From what I understand, the original antennas are what I would call straight 
> preamp with no pre-selection / filtering.  
> 
> The ordered and now inbound replacements are said to contain a SAW filter 
> system. It is the intent of the client to just place these "improved 
> antennas" in 
> to service and get on with life. 
> 
> I would suspect a GPS antenna (and receiver) could be subject to RF overload 
> or blocking, however, we're assuming nothing major has changed at the site, 
> nor 
> any nearby location.  One might think there are more GPS receivers being 
> pushed 
> out of reliable operation by the world around them, I'm just not hearing 
> those stories 
> from a lot of people using them (GPS receivers). 
> 
> Any new install GPS receiver antenna ordered will/should contain some 
> pre-selection 
> to potentially avoid a problem, even some years down the road? Seems like 
> that's 
> where things are going... no more off the shelf, wide band, (hot) 
> preamplified GPS antennas 
> in busy locations? 
> 
> Thank you in advance for any related comments and/or opions ... 
> 
> cheers, 
> 
> skipp 
> 
> skipp025 at jah who dot calm 
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[time-nuts] Re: dual supplies Re: Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

I seem to recall TI having similar parts.

The big gotcha is tossing large chunks of C onto the ground to rail 
connections. A typical op amp is not at all happy with this. 

Bob

> On Jul 10, 2022, at 5:48 PM, Richard (Rick) Karlquist via time-nuts 
>  wrote:
> 
> 
> 
> On 7/10/2022 10:14 AM, Lux, Jim via time-nuts wrote:
>> Yeah, but that virtual ground brings with it it's own set of problems. For 
>> instance, it has to both sink and source current, so you can't just use a 3 
>> terminal regulator to create the midpoint, although I've seen schemes with a 
>> resistor from virtual ground to negative supply, but that's not very power 
>> efficient - the resistor needs to see, say, 10x the maximum sink current.
> 
> You might look at the LT1118-2.85 "Supply Splitter".  It is able to do
> either sink or source, as needed, unlike an ordinary regulator.
> 
> Rick N6RK
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[time-nuts] Re: What about the frequency discrimination method? (offshoot from DIY PN analyzer)

[Bob kb8tq via time-nuts]

Hi

If you dig into the various books on phase noise, they do go into other 
ways to measure it. The bottom line is that things like frequency discriminators
are quite noisy (floor wise) compared to a single mixer. 

Bob

> On Jul 10, 2022, at 10:26 AM, ed breya via time-nuts 
>  wrote:
> 
> Hi Magnus,
> 
> I know what you mean about not needing a quadrature splitter - if you have a 
> very wide phase or delay tuning range - but I'm picturing getting most of the 
> way to quadrature with a fixed structure for a given frequency, and only 
> fine-tuning the phase over a narrow range, in order to minimize the PLL's 
> overall noise contribution. This should also keep it monotonic - too wide a 
> range may let it get stuck on the humps.
> 
> For amplitude calibration, I'm picturing rearranging the splitter ports or 
> guts somehow (as simply as possible) to present the DUT signal to the mixer 
> at 0 or 180 degrees, which should give a maximum DC out.
> 
> BTW I had never heard of the Tayloe detector, but it appears to be the method 
> (4x f multiplier then digital quadrature divide) used in lock-in analyzers, 
> and I have used the same in a number of projects.
> 
> 
> Azelio wrote:
> "How can you measure something, any type of measure, not only PN,
> without a reference? Voltmeters need voltage references, "timemeters"
> (and frequency meters) need time references."
> 
> Azelio, this is a well known technique - I haven't described anything new, 
> just a particular implementation I've been pondering.
> 
> Ed
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

There’s really no need to use the Vref out of the OCXO at all. 
Since many devices don’t have one, you will need a “replacement” 
at some point. Simply pulling the “set reference” off of a cleaned
up output of your main supply(s) is typically how it is done. 

The most basic reason to not hard wire a specific device into the
circuit is to allow a A to B / B to C / C to A swap process to be done. 
That is about the only way to get close to working out the numbers
on this or that reference. Without that data, you are flying blind as
you get close to the limits of the reference. 

Given the characteristics of the mixer and the other stuff involved,
with a roughly +7 dbm input, anything past -174 dbc / Hz is suspect. 
-180 dbc / Hz is significantly better than what you likely can do with
this approach. Indeed, it also is quite a bit better than what you will
find your signal sources doing so that’s not a major constraint. 

Yes there is a wonderful “bet a beer” / after work conversation to be
had about the ultimate phase noise of a +7 dbm signal. More or less,
the bet is won by postulating a 1 ohm source impedance. For real 
world sources …. not so much ….

Bob

> On Jul 10, 2022, at 7:52 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> I've updated the schematic to include the latest additions and added some new 
> measurements
> 
> Schematic: http://athome.kaashoek.com/time-nuts/PNA/Simple_PNA.pdf
> 
> The resistor values (many 18k) are a bit weird but I happen to have a big box 
> of 18k resistors.
> The value of the low pas filter after the mixer (C2,C3,L1) are probably 
> wrong. Calculate yourself for the corner frequency you want.
> The elco's in the PI_controller and the input of the Audio_LNA are probably 
> going to explode due to reverse polarity.
> The output of the REF_Buffer acts as the virtual ground so care was taken 
> (almost) not to draw any current, except for the input of the Audio_LNA.
> The supply of the opamps is not drawn but its from Ground and Vcc (+12V)
> I've tested symmetric supply but the combination of the REF output voltage 
> from the DOCXO and the REF_Buffer provided the least noise.
> The audio_LNA has a gain of 1 for DC and increasing to 100 for for 1Hz and 
> above
> The R/C values around the PI_Controller have not been optimized but they work.
> As the Summer OPAMP inverts to 5-10V the Inverter OPAMP brings it back to 
> 0-5V for the Vtune of the DOCXO
> The LED's provide visual feedback on the tuning. IF both are just on the PLL 
> is in lock. It may be better to have two LED's in series at each side to 
> increase the dimming.
> 
> Some measurements.:
> All indicated levels are 40dBc/Hz higher compared to actual.
> The noise floor: http://athome.kaashoek.com/time-nuts/PNA/PN_baseline_3.JPG
> This is measured without DUT input.
> 
> Rigol signal generator generating 10MHz Phase modulated with 60 degrees noise 
> at -80dBc/Hz: http://athome.kaashoek.com/time-nuts/PNA/
> 
> Rigol signal generator generating 10MHz phase modulated with 0.006 degrees at 
> 220Hz : http://athome.kaashoek.com/time-nuts/PNA/PN_Rigol_3_0.006.JPG
> The 220Hz is under the cursor at -27dBc, at 0.006 degrees modulation it 
> should be at -88dBc, so there must still be a big mistake somewhere.
> 
> AR60 Rubidium reference: http://athome.kaashoek.com/time-nuts/PNA/PN_Rb_3.JPG
> All seems OK, a bit of 50Hz and harmonics.
> 
> OCXO : http://athome.kaashoek.com/time-nuts/PNA/PN_OCXO_3.JPG
> very weird spurs between 40 and 50 Hz
> 
> The famous cheap Chines TCXO: 
> http://athome.kaashoek.com/time-nuts/PNA/PN_TCXO_3.JPG
> Not too bad for offsets of 100Hz and higher but at 10Hz and lower its 20dB 
> worse.
> 
> A home designed/build arduino GPSDO: 
> http://athome.kaashoek.com/time-nuts/PNA/PN_GPSDO_3.JPG
> The GPSDO has a good ADEV but is clearly very noisy!
> 
> I also measured a Marconi 2022 signal generator and it was possible to lock 
> but the phase noise was terrible with strong factional PLL spurs.
> I also tried to measure the phase noise of an old Philips analog 10Hz to 
> 12MHz signal generator but it was impossible to get a lock because the 
> generator output is jumping around several Hz at 10MHz output.
> 
> The noise floor of the simple PNA leaves a lot to improve (from -140dBc/Hz at 
> 10kHz to -180dBc/Hz with better OCXO, LNA and correlation) but it proved to 
> be able to do a first assessment of some not too good oscillator performance.
> 
> Feedback welcome as these are my first baby steps on phase noise nuttery.
> Erik.
> 
> 
> 
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

Pretty much the best mixer to use for this in a basement / DIY basis
is a Mini Circuits RPD-1 or one of it’s siblings. It has a 500 ohm
output on the mix port instead of 50 ohms. Yes, you open circuit
terminate it ( so 5K  ) but as noted, it’s the Zout of the mixer that likely
sets what the op amp sees. With it’s higher output impedance, you 
are even less driven to nutty low noise op amps and 4 ohm feedback
resistors. The good old OP-27 / OP-37 sitting in the dusty back of your
parts drawer from back in 1993 will do just fine. 

Yes, this all gets back to being nutty as you get close to carrier. If you
are after -150 dbc / Hz at 1 Hz offset, you will need go a bit crazy. If
you head this way, there are a lot of posts back in the archives leading
you down various paths to get it done. 

While others have indeed fried expensive setups while loosing a supply
leg, I’ve never run into that problem. It most certainly can happen. I’ve
never taken any special precautions and have yet to “get bit” by the issue. 
As a rough guess, I’d say I’ve powered up various implementations these 
setups a couple thousand times over the years. 

Bob

> On Jul 10, 2022, at 6:32 AM, Gerhard Hoffmann via time-nuts 
>  wrote:
> 
> Am 2022-07-09 22:06, schrieb Erik Kaashoek via time-nuts:
> 
>> Ultra low noise opamps have been ordered to hopefully reduce the internal
>> noise of the PNA but the reference OCXO may already be the limiting factor.
>> The REF voltage output of the OCXO turned out to be rather clean. Much
>> cleaner than a 7805 voltage regulator
> 
> The existence of my own ultra-low noise amplifiers was originally triggered
> by this problem but has turned into a sport of it's own. Don't yield to the
> temptation of driving this too far. A single AD797, LT1028, or ADA4898-2
> all deliver an input noise density of abt. 1nV/rtHz which is the thermal
> noise of a 60 Ohm resistor. ADA4898 has goof price/performnce.
> 
> The diodes in the mixer can easily feature RS = 20 Ohms, and the 2 conducting
> diodes then show 40 Ohms, which is not much less than the 60 Ohm equiv of the 
> opamps.
> RS is ohmic resistance of silicon and contacts, not the differential
> slope resistance of the diode which is only half-thermal IIRC.
> 
> High level mixers often have additional resistors in series to the diodes.
> It's no wonder then that high level mixers are usually not the winners in
> dynamic range. Maybe an array of low-level mixers that are Wilkinsoned
> together on RF and LO, with the IF ports in series would give good results.
> 
> 1. Stephan R. Kurtz, Watkins-Johnson:  Mixers as Phase Detectors
> 2. Bert C. Henderson, W-J: Mixers: Part 2  Theory and Technology
> Copyright © 1981 Watkins-Johnson Company
> Vol. 8 No. 3 May/June 1981
> Revised and reprinted © 2001 WJ Communications, Inc.
> 
> cheers, Gerhard
> 
> 
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

Yes it is a pain to implement dual supplies. I ponder that issue every time 
I build one of these setups. I’ve built a lot of them …. If you are going to 
do a single supply, setting up a “virtual ground” is probably the best way
to go. Do it with a drive circuit to provide very clean 15V off of a 30V supply 
then tack everything ( including *all* the mixer grounds to that 15V supply. 

Keeping the signal undistorted before you check the beat note and use it
to drive the EFC does keep you out of various issues. You do not want to
deal with possible clipping / saturation artifacts getting into either process. 

With devices having positive side EFC, negative side EFC, and “both
sides” EFC, it’s hard to get around a dual supply in any sort of general 
purpose device. A 15V center point is not going to fit any EFC that I’ve
seen :). 

Struggling with the ground loop problem is always the big deal in any
setup. Trying to rule out / take out line noise is usually the final straw in
any series of tests. Doing that with everything at “real ground” is just a 
bit easier. 

Part of the calibration is measuring the beat note as it goes past zero. 
The ’scope gets cranked up and you look at a bit of the crossing right
at ground. Keeping the device happy while doing this is much easier if
the chassis does not need to float at 15V. 

Whatever is used as a supply turns out to be a dedicated device. The 
same ground loop / isolation stuff get in here. An old style non-switching
design is just about mandatory. Keeping switching artifacts out of things
is almost impossible. All of this makes a “build from scratch” approach
less and less crazy. Old style three terminal regulators ( so 78x18 / 79x18 )
are not as easy to find these days. They do fine if you happen to have a 
pair …. There’s really not much power used by any of this. The need for 
anything massive. 100 ma out of each side is overkill ….

As you build things up, you eventually come to the realization that a big
sheet of brass is a good idea for the ground. Tie this and that to the sheet. 
Keep everything non-essential away and likely keep it turned off. Tying
a dedicated supply to that sheet along with the amp and EFC stuff is not
at all unusual. 

Bob

> On Jul 9, 2022, at 11:11 PM, Erik Kaashoek  wrote:
> 
> Hi Magnus,
> Yes, and it works very well, locking is easier as once locked it nicely 
> stay's in lock, , even with a slow drift of either the DUT or the reference. 
> As I could not find a bipolar capacitor the tuning potmeter has to be kept at 
> the low side to avoid blowing the integration capacitor. Maybe a back to back 
> series capacitor with pull down resistor is safer to use.
> Will need to update the schematic to show the small improvements.
> 
> @Bob,
> You mentioned "dual supplies with high voltage" for the first gain opamp. How 
> much impact would dual voltage bring as its a pain to implement.
> I understand everything gets ground reference and you loose the noise of the 
> buffer opamp but as the first gain opamp is in differential mode for its 
> input it does not see the noise of the buffer opamp. Or am I making a mistake?
> 
> On 10-7-2022 2:02, Magnus Danielson via time-nuts wrote:
>> Have you attempted doing a PI-loop as I've suggested? 
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

(see below)

> On Jul 9, 2022, at 12:06 PM, Erik Kaashoek  wrote:
> 
> Getting the simple PNA to lock was a bit difficult due to the overly 
> simplistic translation of the mixer output to the Vtune of the OCXO
> To get some more flexibility I added a summing opamp that summed the mixer 
> output with the output of the coarse tuning potmeter. As the summing causes 
> inversion one extra inverting opamp was added. This made the loop gain 
> constant

Putting some sort of “gain switch” on the summing amp can help in getting the 
loop gain to the point it is usable. 

> To ensure the mixer is in quadrature another opamp was added that amplified 
> the mixer output into two LEDs. One LED on when below zero ouput from mixer, 
> the other on when above zero and both dim when zero output. This made tuning 
> the coarse frequency simple. Turn till the blinking stops and both LED's 
> light up dim. The fine frequency potmeter was no longer needed and the 
> frequency counter is also no longer needed to get into lock

That sounds right. 

> With the summing opamp it is also possible to add an integrator but this has 
> not been done yet.

Typically a simple roll off cap on the feedback R is about all that is done. 

> Shielding is now the biggest problem as any nearby coax connected to a 10MHz 
> source will cause a huge amount of spurs when not at exactly the same 10MHz

Terminating unused devices “at the socket” is often the only way to keep things 
reasonable.

> Ultra low noise opamps have been ordered to hopefully reduce the internal 
> noise of the PNA but the reference OCXO may already be the limiting factor.

Even a “not so fancy” op amp should do pretty well. The big deal is to get to 
dual supplies 
with a fairly high voltage on the first stage. 

> The REF voltage output of the OCXO turned out to be rather clean. Much 
> cleaner than a 8705 voltage regulator

The Ref voltage likely also supplies the oscillator. It can be 20 to 40 db 
“noisier” than the phase 
detector output and have little or no impact on the oscillator performance. 
Yes, there likely is some
filtering between the regulator and the oscillator ….

Bob

> Erik
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

As noted in another post, the phase detector guys talk
about a figure of merit that is not directly comparable to 
the floor of a DBM. If I translate the -170 dbc/ Hz at 100MHz
on the DBM, to the PLL chip FOM, I would add 80 db. That
would make it a -250 dbc FOM vs the -231. 

Since the FOM stuff does not really apply to a DBM approach,
you would never see this done. One works one way, the other
works in a very different fashion. 

Bob

> On Jul 8, 2022, at 8:19 PM, Mike Monett via time-nuts 
>  wrote:
> 
> To Bob kb8tq. You wrote:
> 
>> Hi
> 
>> The noise floor of the double balanced mixer (used as a phase
>> detector at 100 MHz) is in the -165 go -170 dbc / Hz range. I've
>> used the parts you are talking about. Their floor is *way* higher.
> 
>> Bob
> 
> I stated the MC100EP140 would not match the Hittite HMC984LP4E. It has 
> -231 dBc/Hz noise.
> 
> -231 dBc is *way* lower than -170 dbc. About 60 dB lower.
> 
> You might be interested in trying it. Only $13.25 at Arrow:
> 
> https://octopart.com/search?q=HMC984LP4E
> 
> Thanks,
> 
> Mike
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

HI

> On Jul 8, 2022, at 7:44 PM, Mike Monett via time-nuts 
>  wrote:
> 
> To Erik:
> 
> …….
> Another item that might be of interest is the PFD. The Hittite
> HMC984LP4E has -231 dBc/root(Hz) of noise, which is quite low. The
> datasheet is at
> https://www.analog.com/media/en/technical-documentation/data-sheets/hmc984.pdf
> 
> …...
> 
> If you would like to eliminate the problem of quadrature lock, the
> Hittite HMC984LP4E PFD might be of interest. The -231 dBc/Hz of
> noise is very low and might be hard to reach with a DBM.
> 
> If you are interested in following up on phase-frequency detectors
> to eliminate the narrow lock range of double-balanced mixers, I can
> supply you with a wealth of information on the design, implementation, 
> and testing. Just let me know if this would help.
> 
> Mike
> <2N3906S.PNG><2N3906G.PNG><54E696BA.ASC><54E696BA.PLT>___
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Ok, since this keeps coming up ….

The chip guys rate their phase detectors in a somewhat unique way. It works
for their product so that’s fine. However you can’t just toss out their number
and quickly compare it to another number from an entirely different approach.
You need to do the math.

The -231 dbc / Hz number quoted above is a “normalized to one hertz carrier”
number. They call it a FOM or “Figure of Merit” due to the normalization. Other
data sheets phrase things slightly differently when referring to the same 
number.

The first hint you get that there’s something going on with the > 200 db noise
number is in figures 11,12, and 13 where they show actual performance at
a couple of frequencies. The noise at “phase noise test set” sort of offsets 
isn’t 
making it past 120 dbc / Hz on those plots.

The quick and dirty explanation is that you translate the “FOM” number by 
10 Log F to get the noise at the operating frequency. So, if you are at 100 MHz,
you add 80 db to the magic 231 db. That gets you to -151. That still sounds ok 
….
but … this is the broadband FOM and not the close in number. It gets worse as
you go closer to carrier …. 

So no, that’s not going to beat a typical RPD-1 based setup.

Bob




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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi



> On Jul 8, 2022, at 10:35 AM, Mike Monett via time-nuts 
>  wrote:
> 
> Bob, you wrote:
> 
>> Mike. One concern I have with active components as mixer is noise.
>> For an SA I designed only a passive DB diode mixer had  low enough
>> output noise.  Would a PF detector as being  an  active component,
>> not create more noise as output? Erik
> 
>> Yes, you are correct. The only thing with a low enough noise floor
>> for good  phase noise measurements (via the  quadrature technique)
>> is some sort of mixer. Normal digital phase detectors have  way to
>> high a noise floor.
> 
>> Bob
> 
> You are talking about old technology. Old tecnology PFD's were built with
> discrete circuits and probably suffered from crosstalk, deadband, ground
> bounce, VCC noise, and noisy input oscillator signals.
> 
> Modern PFD's have very low noise. For example, the Hittite HMC984LP4E
> digital phase-frequency detector has -231 dBc/Hz of noise and goes up to
> 350MHz:
> https://www.analog.com/media/en/technical-documentation/data-sheets/hmc984.pdf

Hi

The noise floor of the double balanced mixer (used as a phase detector
at 100 MHz) is in the -165 go -170 dbc / Hz range. I’ve used the parts you
are talking about. Their floor is *way* higher.

Bob

> 
> Too bad the price jumped enormously when Analog bought Hittite.
> 
> The MC100EP140 Phase-Frequency Detector has 200 femtoseconds of jitter and
> goes up to 2GHz. That is not going to match the HMC984LP4E but will be
> adequate in many applications:
> https://www.onsemi.com/pdf/datasheet/mc100ep140-d.pdf
> 
> Modern synthesizer IC's have PFD's as the frequency detector and offer very
> low noise.
> 
> You also forget that double-balanced mixers are also very noisy. For
> example, most receivers need a good low noise preamp in front of the mixer
> to get an acceptable noise figure. I am told that part of the reason for
> the high DBM noise is multiple harmonics are generated by the internal
> signals, which combine as part of the output signal.
> 
> Mike
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

Indeed you can switch the gain of the amp. You still need to provide
a low gain output to feed the EFC input on your reference. The chain
to feed the sound card will be crazy high gain for the typical TCXO or
OCXO. Don’t even think of running that sort of gain into a VCO ….

Bob

> On Jul 8, 2022, at 9:16 AM, Erik Kaashoek  wrote:
> 
> Bob,
> Clear, you have a lot more experience and knowledge. For me this is typical a 
> case of "If you don't know about something it must be simple"
> So best would be to make it possible in the simple PNA to switch off the 
> opamp gain, without changing the impedance the mixer sees,  so the offset 
> tuned signal can be used to calibrate the slope.
> I found this picture very helpful to understand the relation between phase 
> modulation depth and the strength of the side bands
> http://athome.kaashoek.com/time-nuts/PM_Sidebands.JPG
> It shows that below 0.2 radian peak phase modulation you can simplify to 
> narrowband FM as only the 1st sideband has relevant power (certainly for the 
> accuracy I am after)
> The whole presentation including the calculation can be found here:
> http://athome.kaashoek.com/time-nuts/Measuring_phase_modulation.pdf
> Written by Bob Nelson from Keysight.
> Very helpful presentation for people (like me) that are new to all this.
> Erik.
> 
> On 8-7-2022 18:58, Bob kb8tq wrote:
>> Hi
>> 
>> Like it or not, the mixer is a non-linear load. It also has a frequency
>> dependence. Even with “saturation” levels, the slope can and does
>> change. That’s the short list, as you dive into it, things get even more
>> complex in terms of “might be” sort of issues.
>> 
>> How can you be in saturation and have the slope change ( it does sound
>> unreasonable) ? The fundamental is not changing much (so you are
>> in saturation). The harmonics of the fundamental are changing. Since
>> the output is actually a triangle wave with rounded “corners” there are
>> indeed harmonics very much present.
>> 
>> The flat parts of the triangle wave are a “good thing” in this case. It
>> makes the device linear over a bit wider range than a sine wave would
>> provide. This gets you out of all sorts of nutty analysis concerning the
>> noise being “to much” to measure with the device. It also relaxes the
>> needed accuracy of the DC lock part of things. ( = slope of a sine wave
>> changes quickly ….).
>> 
>> You never really get away from the “to much noise” question. The
>> common definition of phase noise is that it’s more than 60 db below
>> carrier. That is really just the commonly used limit for  “you may need
>> to think about FM sidebands”.  Yes, that’s another rabbit hole to wander
>> down ….
>> 
>> Bob
>> 
>>> On Jul 8, 2022, at 8:32 AM, Erik Kaashoek  wrote:
>>> 
>>> Bob
>>> This confuses me.
>>>> The calibration of the system changes ( or can change ) each and every 
>>>> time you swap
>>>> out signal sources. The levels are not going to be consistent setup to 
>>>> setup. Thus you
>>>> calibrate each and every time you change out either device.
>>> Assuming each source is saturating the mixer sufficiently (to be confirmed 
>>> by measuring the output level of the source into 50 ohm) I do not 
>>> understand how changing a source can change the calibration. Can you 
>>> explain what is happening?
>>> Please keep in mind I'm not after 0.1dBc/Hz accuracy, +/- 5dBc/Hz would 
>>> already be great.
>>> Erik.
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

Like it or not, the mixer is a non-linear load. It also has a frequency
dependence. Even with “saturation” levels, the slope can and does 
change. That’s the short list, as you dive into it, things get even more
complex in terms of “might be” sort of issues. 

How can you be in saturation and have the slope change ( it does sound
unreasonable) ? The fundamental is not changing much (so you are 
in saturation). The harmonics of the fundamental are changing. Since
the output is actually a triangle wave with rounded “corners” there are 
indeed harmonics very much present. 

The flat parts of the triangle wave are a “good thing” in this case. It 
makes the device linear over a bit wider range than a sine wave would
provide. This gets you out of all sorts of nutty analysis concerning the
noise being “to much” to measure with the device. It also relaxes the 
needed accuracy of the DC lock part of things. ( = slope of a sine wave
changes quickly ….). 

You never really get away from the “to much noise” question. The
common definition of phase noise is that it’s more than 60 db below
carrier. That is really just the commonly used limit for  “you may need 
to think about FM sidebands”.  Yes, that’s another rabbit hole to wander 
down ….

Bob

> On Jul 8, 2022, at 8:32 AM, Erik Kaashoek  wrote:
> 
> Bob
> This confuses me.
>> The calibration of the system changes ( or can change ) each and every time 
>> you swap
>> out signal sources. The levels are not going to be consistent setup to 
>> setup. Thus you
>> calibrate each and every time you change out either device.
> Assuming each source is saturating the mixer sufficiently (to be confirmed by 
> measuring the output level of the source into 50 ohm) I do not understand how 
> changing a source can change the calibration. Can you explain what is 
> happening?
> Please keep in mind I'm not after 0.1dBc/Hz accuracy, +/- 5dBc/Hz would 
> already be great.
> Erik.
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

(see below)

> On Jul 7, 2022, at 10:10 PM, Erik Kaashoek  wrote:
> 
> Bob,
> You may have explained this before but I still do not understand.
> Does the phase modulation slope at the detector depend on the depth of the 
> phase modulation? I think not.

The “phase modulation” you are looking at when observing the slope with a beat 
note
is a full 2-pi radians of modulation for every cycle of the beat note. Since 
that’s guaranteed
with no further effort, it makes a nice standard to use.  There *is* no 
modulation being done
to either signal in this case. 

> With 57 degrees one should get an output voltage that is to be regarded as 
> the 0dBc level but this can not be measured due to the high gain in the audio 
> path.

Which is why you want a two op amp approach. This also gets you a nice path to 
use
for the DC feed for lock. 

> When you reduce the modulation depth with a factor 10 the measured output 
> voltage should decrease with 20dB.

Except you didn’t start out modulating either signal. You simply unlocked them 
and got 
a result that happens to provide 2 pi radians of signal at the output of the 
mixer. 

> Modern digital signal generators are supposed to provide phase modulation 
> with at least 0.01 degree accuracy.
> So it could be possible to measure the phase detector slope with 0.57 phase 
> modulation depth by measuring what should be -40dBc
> Or, if the gain is very high, less accurate with 0.06 phase modulation.
> Or am I making a mistake in my reasoning?

The calibration of the system changes ( or can change ) each and every time you 
swap
out signal sources. The levels are not going to be consistent setup to setup. 
Thus you
calibrate each and every time you change out either device. 

Since signal generators are not likely to get you to the same sort of noise 
levels as a 
very good stand alone source, you very much do not typically want a signal 
generator 
involved in a real measurement. Yes, there are always exceptions to any blanket 
statement …

Bob

> Erik.
> 
> 
> On 8-7-2022 3:57, Bob kb8tq via time-nuts wrote:
>> Hi
>> 
>> One consideration:
>> 
>> If you do signal injection for calibration, you have the amplitude 
>> uncertainties on
>> both the “carrier” and injected signals. The slope at zero on the beat note 
>> is likely
>> to be *much* more accurate ( even if gain measurement at audio gets thrown 
>> in …)
>> 
>> Bob
>> 
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

One consideration:

If you do signal injection for calibration, you have the amplitude 
uncertainties on 
both the “carrier” and injected signals. The slope at zero on the beat note is 
likely
to be *much* more accurate ( even if gain measurement at audio gets thrown in …)

Bob

> On Jul 7, 2022, at 5:19 PM, Magnus Danielson via time-nuts 
>  wrote:
> 
> Hi,
> 
> A well established method is to use a separate offset RF generator that you 
> can steer frequency to form suitable offset and amplitude to form known 
> level. You can now inject this ontop of a signal to measure. Consider that 
> you steer your offset frequency to be +1 kHz of the carrier you measure, and 
> you set the amplitude to be -57 dB from the carrier. This now becomes 
> equivalent to having a -60 dBc phase modulation at 1 kHz.
> 
> The RF generator does not have to be ultra-clean in phase noise just 
> reasonably steerable in frequency and amplitude.
> 
> Cheers,
> Magnus
> 
> On 2022-07-07 12:47, Erik Kaashoek via time-nuts wrote:
>> Bob, others.
>> It has been explained that for the best phase noise level calibration on 
>> should use a signal with one radian phase modulation and measure the output 
>> voltage.
>> The problem with this approach is the unknown gain of the path into the PC. 
>> And due to the gain one can not modulate with one radian as this saturates 
>> the whole path.
>> An alternative method for phase noise level calibration could be to create 
>> an oscillator so bad its phase noise can be measured using a spectrum 
>> analyzer. To make such a bad oscillator a 10MHz signal was phase modulated 
>> with noise. The phase noise became visible on the spectrum analyzer just 
>> above 20 degrees of modulation. The phase noise level saturated between 55 
>> and 60 degrees which is consistent with one radian (57 degrees). The 
>> spectrum analyzer could measure the phase noise at a flat -80dbc/Hz ( yes 
>> Bob, I better use the right dimensions)
>> The simple phase noise analyzer also measured the phase noise at -80dBc 
>> providing evidence the level calibration was done correctly.
>> I also tried to increase the DUT drive into the mixer further above 
>> saturation so see if this made any change in the measured level but once 
>> above 0dBm I did not observe any change up to +10dBm drive. Any higher 
>> levels felt too dangerous.
>> There is still a lot of work to be done to further increase accuracy.
>> Erik.
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

The idea is partly to lock the two devices. The bigger objective is to hold
the mixer output at the correct zero volt operating point. 

Cabling things to a different device and then doing phase correction to keep
things at zero would be a major pain. 

Bob

> On Jul 7, 2022, at 5:52 AM, Mike Monett via time-nuts 
>  wrote:
> 
> You wrote:
> 
>> Mike.
>> One concern I have with active components as mixer is noise. For an SA I
>> designed only a passive DB diode mixer had low enough output noise. Would a
>> PF detector as being an active component, not create more noise as output?
>> Erik
> 
> Eric, you do not have to give up your double balanced mixer. You can use a
> phase/frequency detector to lock the reference to the DUT, and still use
> the DBM to do the phase noise analysis.
> 
> Here is a block diagram of the circuit: pna.png
> 
> I don't know if this is going to work, so I will send this email and wait
> for it to show up in the lists. If it does, I have a lot more information
> for you.
> 
> 
> 
> 
> 
> 
> 
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

Yes, you do need to know the system gain. Since we are talking about
gain at audio, measuring the gain directly is not a crazy thing to do. One
of the things that makes audio spectrum analyzers a nice tool for this that
they eliminate the “variable gain to the sound card” issue. 

Some sound card setups are a lot easier to work with than others. If you
are restricted to the sound input on your motherboard things can get a bit
crazy. It is not unusual for folks to dig up a “pro” (whatever that means 
on a sound card ) card that has better drivers and more access to this and
that.

Given how fast the PC world changes, the board that was a wonderful thing
last time somebody dove in, likely is long out of production by now. The drivers
that made it work so well may have been “improved” and it no longer gives
you the control it once did. This makes for a bit of trial and error to get it 
all
going.

Bob

> On Jul 7, 2022, at 2:47 AM, Erik Kaashoek  wrote:
> 
> Bob, others.
> It has been explained that for the best phase noise level calibration on 
> should use a signal with one radian phase modulation and measure the output 
> voltage.
> The problem with this approach is the unknown gain of the path into the PC. 
> And due to the gain one can not modulate with one radian as this saturates 
> the whole path.
> An alternative method for phase noise level calibration could be to create an 
> oscillator so bad its phase noise can be measured using a spectrum analyzer. 
> To make such a bad oscillator a 10MHz signal was phase modulated with noise. 
> The phase noise became visible on the spectrum analyzer just above 20 degrees 
> of modulation. The phase noise level saturated between 55 and 60 degrees 
> which is consistent with one radian (57 degrees). The spectrum analyzer could 
> measure the phase noise at a flat -80dbc/Hz ( yes Bob, I better use the right 
> dimensions)
> The simple phase noise analyzer also measured the phase noise at -80dBc 
> providing evidence the level calibration was done correctly.
> I also tried to increase the DUT drive into the mixer further above 
> saturation so see if this made any change in the measured level but once 
> above 0dBm I did not observe any change up to +10dBm drive. Any higher levels 
> felt too dangerous.
> There is still a lot of work to be done to further increase accuracy.
> Erik.
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[time-nuts] Re: Isolation amp transistors

[Bob kb8tq via time-nuts]

Hi

> On Jul 7, 2022, at 12:09 AM, Gerhard Hoffmann via time-nuts 
>  wrote:
> 
> Am 2022-07-07 7:22, schrieb Bob kb8tq via time-nuts:
>> Hi
>>> On Jul 6, 2022, at 1:53 PM, Richard Karlquist via time-nuts 
>>>  wrote:
>>> The 2N5179 has high base spreading resistance (decreases isolation).
>> As does sticking a resistor (even a small one) in series with the base
>> …. Yes, inductance is even worse.
> 
> and at frequencies where beads work, they also generate thermal noise,
> like any other dissipative thingie. See the sim of a random ferrite bead
> from the LTspice library. V1 is only there as a compiler pleaser to
> enable the proper syntax. It really has no influence.

A cascode buffer is a very common thing in an oscillator. Over the decades
folks have made a *lot* of them. Having them turn into oscillators at UHF /
microwave frequencies is not at all unusual. Having the loose isolation for 
various reasons ( somebody used the wrong bypass cap maybe …) is also 
not at all unusual. 

Coming up with a model for a spice analysis that will always catch these 
things is non-trivial. A transistor intended to be used to < 50 MHz rarely has
a model that includes everything that’s relevant at 1.8 GHz. 

> 
>> For “best isolation” in a cascode you very much want the base of
>> the common base stage nailed to ground. Typically “lower” Ft transistors
>> with a decent base structure are the best choice for the common base stage.
>> Both stages benefit from low 1/F noise > in the audio range if this is for
>> a phase noise test set.  This is why people use what would normally be
>> considered “audio” transistors ….
> 
> Cascodes do not add much noise when they have a decent beta. Zin at the
> Emitter is a few Ohms and Zout of the driving stage is maybe KOhm. That
> makes the driving stage DICTATE the collector current. Also, 1/f noise
> is not THAT bad since the load resistance is near 0 in the 1/f region.
> Thus, at least no gain at 1/f frequencies. In a linear amplifier, it
> would not get mixed up anyway. Makes me like the Driscoll oscillator.
> 
> I could not find a transformer (Macom, Pulse Eng.) that provided an
> acceptable S22.

In this case, the amplifier is driving into a double balanced mixer that has
a *very* similar transformer. Whatever you are seeing with the part you 
buy off the shelf is already there in the next stage. 

If you are trying for S22 from DC to light then yes, transformers have issues.
They also have issues if your “acceptable” number something past 40 db. 
Mixers (used as phase detectors) need to see termination at fairly specific
frequencies. This helps quite a bit.  

Bob

> The resistive 50 Ohm in par with (4.7u + bead) worked best.
> At least the momentary collector voltage can exceed the supply.
> Appreciated in the light of 12V operation. But without the transformer,
> one pays with a lot of bias current, therefore sot-89.
> The circuit is not exact, it was in the middle of what/if experiments.
> Thus, some funny values.
> The cascode is mostly harmless (TM). What hurts, that is Q2 stability.
> 
> @Florian:  150 mA, not too much Vce required, 1-1.5 GHz ft.
> 
> Cheers, Gerhard
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[time-nuts] Re: Isolation amp transistors

[Bob kb8tq via time-nuts]

Hi

The tube cascode has it’s own issues. Setting up a tube circuit for
the sort of isolation we are talking about here is very difficult. 

Bob

> On Jul 6, 2022, at 9:46 PM, glenlist via time-nuts  
> wrote:
> 
> how about grounded grid ?
> 
> Bob can you get  better isolation with a vaccuum tube cascode than a solid 
> state cascode ?
> 
> -glen
> 
> On 07/07/2022 15:22, Bob kb8tq via time-nuts wrote:
>> Hi
>> 
>>> On Jul 6, 2022, at 1:53 PM, Richard Karlquist via 
>>> time-nuts  wrote:
>>> 
>>> The 2N5179 has high base spreading resistance (decreases isolation).
>> As does sticking a resistor (even a small one) in series with the base …. 
>> Yes, inductance
>> is even worse.
>> 
>> For “best isolation” in a cascode you very much want the base of the common 
>> base
>> stage nailed to ground. Typically “lower” Ft transistors with a decent base 
>> structure
>> are the best choice for the common base stage. Both stages benefit from low 
>> 1/F noise
>> in the audio range if this is for a phase noise test se
>> 
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[time-nuts] Re: Isolation amp transistors

[Bob kb8tq via time-nuts]

Hi

> On Jul 6, 2022, at 1:53 PM, Richard Karlquist via time-nuts 
>  wrote:
> 
> The 2N5179 has high base spreading resistance (decreases isolation).

As does sticking a resistor (even a small one) in series with the base …. Yes, 
inductance
is even worse. 

For “best isolation” in a cascode you very much want the base of the common 
base 
stage nailed to ground. Typically “lower” Ft transistors with a decent base 
structure
are the best choice for the common base stage. Both stages benefit from low 1/F 
noise
in the audio range if this is for a phase noise test set.  This is why people 
use what would
normally be considered “audio” transistors ….

Bob


> 
> ---
> Rick Karlquist
> N6RK 
> 
> On 2022-07-06 12:18, ed breya via time-nuts wrote:
> 
>> My favorite VHF Q is the good old 2N5179 or similar, but it appears you want 
>> something in surface mount, and not obsolete. I'm not familiar with the 
>> modern SMT stuff. If your present transistors are working, but just need a 
>> bit more stability, it seems it should be OK with the right scheme, and not 
>> the transistors' fault.
>> 
>> Ed
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[time-nuts] Re: Should a double oven XO be thermally isolated or just draft protected?

[Bob kb8tq via time-nuts]

Hi

Unless you measure the change of the device over a controlled temperature
range ( like 0 to 70C ) at a controlled rate ( like < 0.1C / minute ) it’s hard 
to 
know if this or that restriction / insulation on an OCXO has “upset” its 
temperature
compensation. If you “make the heater work half as hard” you may have doubled 
the thermal gain. That’s big change ….. 

Bob

> On Jul 5, 2022, at 1:48 PM, ed breya via time-nuts  
> wrote:
> 
> This may give some idea of how fast things can happen when the OCXO is 
> subject to drafts. I have this dual GPSDO box that usually is open for 
> experimenting, and have a setup comparing one of the 10 MHz outs to my 
> portable Rb reference. The 10 GHz multiplied output from the Rb is indicated 
> on a microwave counter, using the GPSDO as reference. This gives 1 mHz 
> resolution on the 10 Mhz signals at the 1 Hz counter resolution limit. It 
> normally reads 10 GHz "exact" +/- 1 Hz when things are stable, or up to maybe 
> up to 2 Hz when garage ambient is changing. I just turn the counter on 
> whenever I'm in the mood to take a look.
> 
> The upper GPSDO board is exposed, so I can just put a finger on the case of 
> the small (about 1" x 1.5") OCXO for a few seconds. Almost immediately, the 
> counter shows several Hz change, which gradually recovers, with some over- 
> and under-shoot. During all this, the OCXO is changing, and the GPSDO is 
> trying to fix it.
> 
> Having a bigger OCXO with more thermal mass and insulation, and having more 
> protection from fast ambient changes can help a lot. As others have said, you 
> don't want to overdo it - the oven heating system must be kept working under 
> all conditions, but it's OK to make it not have to work too hard.
> 
> An extreme example of a bad thermal situation is in the beloved HP8566. I 
> have often lamented about the poor placement of its internal OCXO, which is 
> right in the main air plenum that feeds the fan cooling air to the whole 
> instrument. The OCXO is subject immediately to any change in ambient, and its 
> heater has to work very hard. I'm convinced that this is the cause of most 
> OCXO failures in the 8566. I've had to refurbish a number of these. The 
> typical failure I've encountered is that the foam insulation deteriorates 
> from the high heat flux needed, and the chemicals from the foam cause the 
> oven setpoint adjustment pot wiper contact to fail. An easy way to spot this 
> problem is to gently shake the OCXO - if you can hear and feel the guts 
> clunking around inside, then it's due for repair.
> 
> At an opposite extreme, in my "Z3801A in a HP5065A carcass" project, I 
> substantially isolate the OCXO from ambient. It's already a double-oven 
> style, and I further enclosed it in a mu-metal box (made from a CRT shield). 
> The OCXO is suspended on rubber vibration mounts, inside the box, and has a 
> thin (~1/4") layer of non-woven fiber insulation on all sides between it and 
> the box. The insulation has very little R-value, but suppresses turbulence 
> and convection flow inside. The Z3801A guts are arranged specially to fit and 
> occupy about two thirds of the cabinet volume, and this section is largely 
> sealed off from the outside and from the right side battery compartment. A 
> small fan runs at very low speed to gently circulate the air inside the 
> compartment, and the plentiful amount of cabinet skin easily dissipates the 
> total power. The same type of insulation is also placed under and atop the 
> main board in the DAC/EFC circuit area, to slow down thermal changes there. 
> The EFC's SMB connector set will also be shrouded with an insulating tube, to 
> reduce thermal voltage. I even changed the nearest board mounting post to 
> plastic, to reduce effects of thermal conduction and ground current in the 
> vicinity.
> 
> All of this does not protect from ambient, but only the rate of change. It's 
> more or less a constant temperature rise type deal, assuming constant power 
> dissipation when everything's stable - and not too much wind or draftiness on 
> the whole cabinet.
> 
> Ed
> 
> 
> 
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer (Erik Kaashoek)

[Bob kb8tq via time-nuts]

Hi

> On Jul 5, 2022, at 9:00 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Mike.
> One concern I have with active components as mixer is noise. For an SA I
> designed only a passive DB diode mixer had low enough output noise. Would a
> PF detector as being an active component, not create more noise as output?
> Erik

Yes, you are correct. The only thing with a low enough noise floor for good
phase noise measurements (via the quadrature technique) is some sort of mixer.
Normal digital phase detectors have way to high a noise floor.

Bob

> 
> On Tue, Jul 5, 2022, 18:20 Mike Monett via time-nuts <
> time-nuts@lists.febo.com> wrote:
> 
>> You stated:
>> 
>> Mike,
>> The phase detector is an ADE-1 mixer, the IF output of the mixer goes
>> into a loop filter that has a corner frequency of about 0.2Hz to enable
>> Phase noise measurements down to 1Hz offset
>> 
>> That is your problem. A double balanced mixer is an exclusive-or phase
>> detector. The lock range is determined by the loop bandwidth, as you have
>> found.
>> 
>> The phase-frequency detector is completely different. It will lock to any
>> signal in the lock range, independent of loop bandwidth. You can have a
>> bandwidth of 0.001 Hz, and it will still lock. Think of what this could do
>> for your phase measurements.
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer

[Bob kb8tq via time-nuts]

Hi

The mixer sensitivity is based on both ports being saturated. You 
are trying to check the circuit end to end in it’s “as used” configuration. 
There is no real need for an added calibration source, provided you 
pay attention to the basics. 

Bob

> On Jul 5, 2022, at 12:04 PM, Erik Kaashoek  wrote:
> 
> Why would using a calibrated noise source into a 1Hz per bin FFT not be a 
> good level check? Just subtract 3 dB due to both widebands folding and you 
> have a reference. 
> A noise source should also be great for checking flatness I think.
> A noise source at -90dBm/Hz around 10 MHz is cheap on eBay. Calibration can 
> be done with any SA having noise markers.
> 
> On Tue, Jul 5, 2022, 21:34 Bob kb8tq mailto:kb...@n1k.org>> 
> wrote:
> Hi
> 
> One of the “gotcha’s” with this is the need for dual supplies. That’s what 
> lets you
> get to an 30 something volt peak to peak output on the initial amp. It also 
> keeps
> all the signals ground referenced through the system. 
> 
> The downside ( as noted in posts from a number of years ago) is that 
> sequencing
> of the supplies can matter. Having just one supply running and the other 
> missing in 
> action can put a bit of current where you might not want it to go. The common 
> example is the mixer output port. Depending on how things are done, there are 
> other possible paths. There are a lot of ways to deal with this. 
> 
> One quick check of “how am I doing?”: Look at the fundamental amplitude of 
> the 
> undistorted beat note. Then look at the 1 Hz normalized noise with both ports 
> driven 
> from the same device. Your noise should be more than 145 db down. The same 
> should also be true of the preamp output with the input terminated in your 
> magic 
> 50 or 500 ohm resistance. Since the preamp likely does not pass the beat note 
> without saturating, there is a bit of math here or there to get this all 
> done. 
> 
> Is this gizmo going to have a flat floor right into 1 Hz? Nope. All these 
> magic floor 
> numbers are talking about >= 1 KHz off carrier. 1/F noise is a very real 
> thing in 
> your mixer, and in the op amp used for the preamp. Fortunately it also 
> impacts 
> signal sources. You are not likely to need -155 dbc / Hz data at 1Hz off 
> carrier. 
> 
> Bob
> 
> > On Jul 5, 2022, at 10:51 AM, Erik Kaashoek  > <mailto:e...@kaashoek.com>> wrote:
> > 
> > Bob, See below
> > Thanks.
> > Erik.
> > 
> > On 5-7-2022 19:37, Bob kb8tq wrote:
> >> Hi
> >> 
> >> One “cute trick” that can be done on the lowpass filter:
> >> 
> >> Ideally you would like to terminate the mixer properly at the DUT and DUT 
> >> x 2
> >> frequencies. Open circuit (or short circuit) is ok at audio. The quick and 
> >> easy way
> >> to do this is to put an appropriate resistor ( 50 ohm, 500 ohm, whatever ) 
> >> in series
> >> with C2 on your schematic. It is common to put an RF choke across it to 
> >> get it
> >> “out of the way” as you head to audio.
> > That is exactly what I did, forgot to show in schematic.
> >> 
> >> Spice is your friend in this case. It’s way easier to pump it into LTSpice 
> >> or something
> >> similar and tinker then trying to come up with some sort of math solution.
> >> 
> >> Stuffing in a resistor at the output of the lowpass is also a common 
> >> thing. Again the
> >> value is a “that depends” sort of thing. 500 ohms or 5K are the likely 
> >> candidates. The
> >> main value is to take the port to zero when the mixer is not being driven.
> > I used 5k
> >> 
> >> No, none of this is a big deal.
> >> 
> >> Since the ref out of the OCXO is likely a pretty noisy item, I would not 
> >> get it anywhere
> >> near the low noise audio part of the circuit. The mixer output is 
> >> typically grounded and
> >> the ref out is ignored.
> > There is an RC filter between the REF out and the buffer opamp, but I can 
> > move the mixer to ground and feed the bottom of the tune potentiometer 
> > (R3/R4), will test, but U2 still need 5V as reference for operation.
> > An easier method may be to connect the auto bias capacitor C1 not to ground 
> > but to the buffered Vref, this puts the opamp into balanced input mode thus 
> > eliminating any remaining noise from Vref.
> >> The EFC is driven off of the lock section of the circuit to keep
> >> things in quadrature.
> >> 
> >> If things are “locking up” without a lock circuit then indeed the earlier 
> >> post abo

[time-nuts] Re: DIY Low offset Phase Noise Analyzer

[Bob kb8tq via time-nuts]

Hi

One “cute trick” that can be done on the lowpass filter:

Ideally you would like to terminate the mixer properly at the DUT and DUT x 2 
frequencies. Open circuit (or short circuit) is ok at audio. The quick and easy 
way
to do this is to put an appropriate resistor ( 50 ohm, 500 ohm, whatever ) in 
series
with C2 on your schematic. It is common to put an RF choke across it to get it 
“out of the way” as you head to audio. 

Spice is your friend in this case. It’s way easier to pump it into LTSpice or 
something 
similar and tinker then trying to come up with some sort of math solution. 

Stuffing in a resistor at the output of the lowpass is also a common thing. 
Again the 
value is a “that depends” sort of thing. 500 ohms or 5K are the likely 
candidates. The
main value is to take the port to zero when the mixer is not being driven. 

No, none of this is a big deal. 

Since the ref out of the OCXO is likely a pretty noisy item, I would not get it 
anywhere 
near the low noise audio part of the circuit. The mixer output is typically 
grounded and
the ref out is ignored. The EFC is driven off of the lock section of the 
circuit to keep 
things in quadrature. 

If things are “locking up” without a lock circuit then indeed the earlier post 
about injection
locking applies.

Bob

> On Jul 5, 2022, at 3:15 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Here a small schematic of the simple Phase Noise Analyzer
> Hope this answers some of the questions.
> The +5 V reference from the OCXO is buffered by U1.
> The 3 ports of the used ADE-1 mixer are galvanically  isolated greatly 
> helping to reduce ground loops.
> The output of the mixer is low passed using C2,L1 and C3 and used as input to 
> the tuning of the OCXO
> R4/R4 are actually two potmeters linked with summing resistors for 
> coarse/fine frequency adjustment.
> Inside the OCXO is a R/C low pass filter with a corner frequency of about 0.5 
> Hz.
> The potmeter setting do influence the loop gain but in practice this is not a 
> problem.
> The mixer output is also amplified by U2 using automatic bias done with a 
> large C1 and send into the audio input of a PC running the FFT program.
> The 10MHz output from the DUT goes into RF_GND and RF_IN
> For simplicity the supply decoupling capacitors and the output DC blocking 
> capacitor are not drawn.
> The opamps use single +12 V, just like the OCXO.
> The OCXO must have better (or just equal) phase noise performance compared to 
> the oscillator being measured (the DUT)
> The +12 V supply comes from a bench supply with floating ground
> 
> By shorting C3 one can check for unwanted behavior like injection locking and 
> measure the  internally generated noise.
> R5 is added to measure noise levels when no DUT is connected.
> 
> The performance is surprisingly good, although one has to use a frequency 
> counter to bring the DUT and OCXO close enough for lock.
> To check for 90 degrees lock the R3/R4 potmeters are tuned to maximum noise 
> level while still having lock.
> Listening to the audio out is like listening to a DSB receiver. One can hear 
> any disturbance or stray 10MHz. like the 10MHz house clock distribution cable 
> being too close (not connected) to the Phase Noise Analyzer.
> Shielding is important to keep the noise down.
> By ensuring the FFT has a bandwidth of 1Hz a calibrated noise source can be 
> use to establish a power level reference, much needed because of the 
> undefined gains in the PC audio path.
> 
> I know this design is far away from what many people on this list are used 
> to, but it was good enough for me to quickly check the performance of some 
> oscillators.
> Erik.___
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer

[Bob kb8tq via time-nuts]

Hi

One of the “gotcha’s” with this is the need for dual supplies. That’s what lets 
you
get to an 30 something volt peak to peak output on the initial amp. It also 
keeps
all the signals ground referenced through the system. 

The downside ( as noted in posts from a number of years ago) is that sequencing
of the supplies can matter. Having just one supply running and the other 
missing in 
action can put a bit of current where you might not want it to go. The common 
example is the mixer output port. Depending on how things are done, there are 
other possible paths. There are a lot of ways to deal with this. 

One quick check of “how am I doing?”: Look at the fundamental amplitude of the 
undistorted beat note. Then look at the 1 Hz normalized noise with both ports 
driven 
from the same device. Your noise should be more than 145 db down. The same 
should also be true of the preamp output with the input terminated in your 
magic 
50 or 500 ohm resistance. Since the preamp likely does not pass the beat note 
without saturating, there is a bit of math here or there to get this all done. 

Is this gizmo going to have a flat floor right into 1 Hz? Nope. All these magic 
floor 
numbers are talking about >= 1 KHz off carrier. 1/F noise is a very real thing 
in 
your mixer, and in the op amp used for the preamp. Fortunately it also impacts 
signal sources. You are not likely to need -155 dbc / Hz data at 1Hz off 
carrier. 

Bob

> On Jul 5, 2022, at 10:51 AM, Erik Kaashoek  wrote:
> 
> Bob, See below
> Thanks.
> Erik.
> 
> On 5-7-2022 19:37, Bob kb8tq wrote:
>> Hi
>> 
>> One “cute trick” that can be done on the lowpass filter:
>> 
>> Ideally you would like to terminate the mixer properly at the DUT and DUT x 2
>> frequencies. Open circuit (or short circuit) is ok at audio. The quick and 
>> easy way
>> to do this is to put an appropriate resistor ( 50 ohm, 500 ohm, whatever ) 
>> in series
>> with C2 on your schematic. It is common to put an RF choke across it to get 
>> it
>> “out of the way” as you head to audio.
> That is exactly what I did, forgot to show in schematic.
>> 
>> Spice is your friend in this case. It’s way easier to pump it into LTSpice 
>> or something
>> similar and tinker then trying to come up with some sort of math solution.
>> 
>> Stuffing in a resistor at the output of the lowpass is also a common thing. 
>> Again the
>> value is a “that depends” sort of thing. 500 ohms or 5K are the likely 
>> candidates. The
>> main value is to take the port to zero when the mixer is not being driven.
> I used 5k
>> 
>> No, none of this is a big deal.
>> 
>> Since the ref out of the OCXO is likely a pretty noisy item, I would not get 
>> it anywhere
>> near the low noise audio part of the circuit. The mixer output is typically 
>> grounded and
>> the ref out is ignored.
> There is an RC filter between the REF out and the buffer opamp, but I can 
> move the mixer to ground and feed the bottom of the tune potentiometer 
> (R3/R4), will test, but U2 still need 5V as reference for operation.
> An easier method may be to connect the auto bias capacitor C1 not to ground 
> but to the buffered Vref, this puts the opamp into balanced input mode thus 
> eliminating any remaining noise from Vref.
>> The EFC is driven off of the lock section of the circuit to keep
>> things in quadrature.
>> 
>> If things are “locking up” without a lock circuit then indeed the earlier 
>> post about injection
>> locking applies.
> Will test for locking with mixer output shorted.
>> 
>> Bob
>> 
>>> On Jul 5, 2022, at 3:15 AM, Erik Kaashoek via time-nuts 
>>>  wrote:
>>> 
>>> Here a small schematic of the simple Phase Noise Analyzer
>>> Hope this answers some of the questions.
>>> The +5 V reference from the OCXO is buffered by U1.
>>> The 3 ports of the used ADE-1 mixer are galvanically  isolated greatly 
>>> helping to reduce ground loops.
>>> The output of the mixer is low passed using C2,L1 and C3 and used as input 
>>> to the tuning of the OCXO
>>> R4/R4 are actually two potmeters linked with summing resistors for 
>>> coarse/fine frequency adjustment.
>>> Inside the OCXO is a R/C low pass filter with a corner frequency of about 
>>> 0.5 Hz.
>>> The potmeter setting do influence the loop gain but in practice this is not 
>>> a problem.
>>> The mixer output is also amplified by U2 using automatic bias done with a 
>>> large C1 and send into the audio input of a PC running the FFT program.
>>> The 10MHz output from the DUT goes into RF_GND and RF_IN
>>> For simplicity the 

[time-nuts] Re: DIY Low offset Phase Noise Analyzer

[Bob kb8tq via time-nuts]

Hi

If you need that sort of isolation, it certainly can be done. 
NIST has papers on very simple / DIY compatible cascode
amps that will do the trick. ( chain of common base stages
driven by a common emitter). Some folks on the list have 
gone a lot further in terms of complexity than NIST did. 

Device wise, the cascode amps seem to work pretty well
with some very humble transistors ( 2N3904 etc ). There
likely are fancier parts out there, but some of the really
old stuff appears to be “good enough”. 

Why cascode ( or just common base) ? You can manage
the gain pretty well. The amps likely are going to be quite
low gain along with low noise. Thats not a great combination. 

There are a lot of OCXO’s out there that don’t need anything
as crazy as the multi stage NIST amps. If you are looking at
a system (as opposed to an oscillator) isolation already has
been addressed in the instrument. 

Part of the “look at the beat note” process is to observe
that it looks like a proper clipped triangle wave. The slopes
on the rising and falling edges should be similar and of a 
reasonable slope. 

If you have injection locking ( and with a typical low frequency
note) the rising and falling edges will begin to distort. The
“lock” process will start to occur as the note crosses zero. 
This is yet another good reason to use a low frequency 
note. It also is good to look at the slope of *both* zero crossings. 

Bob

> On Jul 5, 2022, at 12:16 AM, Leon Pavlovic via time-nuts 
>  wrote:
> 
> I've tried to do a simple and low-cost sound-card PN system too back in the
> days. It's fun, when you see the broadband thermal resistor noise in your
> FFT plots :) But when doing experiments with the crystal oscillators, I've
> found out that one of the *MOST* important things to worry about is
> injection locking, especially if you do OCXOs and very low PN oscillators.
> 
> Before feeding the RF and LO ports of the mixer, you should have isolation
> amps on both ports with about or more than 100dB S12 isolation, not to get
> into trouble. Without them, you'll be measuring unreal PN noise...
> 
> Cheers,
> Leon
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[time-nuts] Re: Silicom PCIe timestamping network cards

[Bob kb8tq via time-nuts]

Hi

> On Jul 4, 2022, at 10:04 AM, Poul-Henning Kamp via time-nuts 
>  wrote:
> 
> 
> Poul-Henning Kamp via time-nuts writes:
> 
>> The timestamping counter gets its clock from the ethernet line
>> signals, and the counting frequency therefore depends on the ethernet
>> speed:
>> 
>>  100 Mb/s1.5625 MHz
>>  1 Gb/s  15.625 MHz
>>  10 Gb/s 156.25 MHz
>> 
>> (The 8ns timestamping mentioned must be something outside the 82599)
> 
> I should probably expand on this to prevent misunderstandings:
> 
> The 82599 chip will timestamp with 6.4ns resolution, and since both
> the frequency and the timestamp edge is derived from the ethernet
> signal when you time packets, there is no noise process involved,
> and you do get your full 6.4ns worth.
> 
> I understand the "8ns" number in the datasheet for the card to refer
> to the PPS input and assume the extra 1.5ns to be noise in the
> analog domain outside the i82599 chip.

Could be. They also mention a 25 MHz clock on the card. That could
get you to a 125 MHz time base with a 8 ns resolution. Again, without
a deep dive into what they did … who knows.

Bob

> 
> -- 
> Poul-Henning Kamp   | UNIX since Zilog Zeus 3.20
> p...@freebsd.org | TCP/IP since RFC 956
> FreeBSD committer   | BSD since 4.3-tahoe
> Never attribute to malice what can adequately be explained by incompetence.
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[time-nuts] Re: Silicom PCIe timestamping network cards

[Bob kb8tq via time-nuts]

Hi

I think the key parameter is the 8 ns resolution on the time stamp. 
That may or may not be adequate for this or that application. 

Without doing a deep dive on the part, it’s not real clear how they
deal with the accuracy of the onboard timebase. It’s rated at 0.01 ppm
with no real details. Obviously if it is free running and there is no
practical calibration approach …. that will add to the error as well.
One would *hope* they thought about this ….you never know...

Bob

> On Jul 4, 2022, at 8:05 AM, John Miller via time-nuts 
>  wrote:
> 
> Hey all,
> I'm curious if anyone here knows much about these silicom timestamping 
> network interfaces? They pop up fairly often for sale, usually for not that 
> much. They have SMA in/out for a PPS signal in addition to the network 
> interfaces. I've found datasheets like this (linked below), but no manuals or 
> software. Has anyone used one of these in a lab before? I'm very tempted.
> 
> https://www.silicom-usa.com/wp-content/uploads/2016/08/PE310G2TSI9P-10G-Precision-Time-Stamping-Server-Adapter.pdf
>  
> 
> 
> Thanks,
> John
> KC1QLN
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[time-nuts] Re: DIY Low offset Phase Noise Analyzer

[Bob kb8tq via time-nuts]

Hi

If you are running a high gain op-amp to buffer things into a
sound card *and* using the same op-amp output to drive the
EFC, then you will have problems. 

Simple answer is to use a couple of op amps.

Buffer the mixer with something low noise. Get the output of the
mixer up to the point it almost saturates the op amp. Just how 
much gain that is depends a lot on your parts and power supplies.
Running +/- 18V supplies into this op amp is no at all unusual.

Since this output is linear, you have the full range of the beat note
present. Nothing has been lost (yet). 

One path off this device goes to the high gain stage to the sound
card. If the beat note is present, you will have clipping there. 

The other path goes to whatever you do to run the EFC. There are
*many* approaches that could be used. One of many is a variable
gain / variable roll off amp to “set” the PLL corner. That is followed
by a simple summing amp to tune out the DC offset on the EFC.

One thing to note is that once you get past the fancy amp in the 
first stage, the noise properties of the other amps are much less
important ( assuming the supplies are fairly high ). This makes the
rest of the circuit pretty cheap to build.

Bob

> On Jul 4, 2022, at 6:02 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Hi David,
> Let me explain the DIY "measurement instrument"
> DOCXO into LO port of a mixer, DUT into RF, IF port with low pass filter to 
> both steer the Vtune of the DOCXO and into an opamp to amplify 1-100kHz into 
> PC audio input. On PC audio spectrum analyzer.
> So yes, the limiting factor is the measurement instrument, probably the opamp
> Further improvements of the setup has reduced the noise level at 10kHz offset 
> to -145dBc/Hz so the phase noise of the DOCXO plus the AR60 is at least 
> -145dBc/Hz at 10kHz offset. If the DOXCO is, as the spec states, -150dBc/Hz 
> at 10kHz offset that the AR60 can not be above -145dBc/Hz at 10kHz.
> 
> On 4-7-2022 12:01, David C. Partridge via time-nuts wrote:
>> Are you sure the PN floor of the measurement instrument isn't the limiting 
>> factor?
>> 
>> David
>> 
>> -Original Message-
>> From: Erik Kaashoek via time-nuts 
>> Sent: 04 July 2022 10:14
>> To: time nuts ; Erik Kaashoek 
>> Cc: Erik Kaashoek 
>> Subject: [time-nuts] DIY Low offset Phase Noise Analyzer
>> 
>> :
>> 
>> At 10kHz offset the Phase Noise of the DOCXO should be -150dBm but
>> unfortunately either the noise of the ultra low noise opamps or the
>> Phase Noise of the AR60 is almost 20dB higher.
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[time-nuts] Re: Should a double oven XO be thermally isolated or just draft protected?

[Bob kb8tq via time-nuts]

Hi

If you tear into *lots* of HP devices with OCXO’s in them (not just the 10811 
version),
the most typical place for the OCXO is right. next to the power supply. That 
puts it inline
with the output of the fan. 

Why? The OCXO gets hot. Heat buildup in the instrument is not a good thing. 
They put
it there to get the heat out of the box as quickly as they can. Since the 
“draft” is a constant
(and not a puff puff puff) it’s not as big a deal as you might think. What it 
does do is to
move the changes in the *outside* environment over to the OCXO more quickly.  
If there
is an impact, it is from your lab ….

Bob

> On Jul 1, 2022, at 12:31 PM, Dr. David Kirkby via time-nuts 
>  wrote:
> 
> On Fri, 1 Jul 2022 at 20:11, Erik Kaashoek via time-nuts <
> time-nuts@lists.febo.com> wrote:
> 
>> I'm trying to build a stable reference for a phase noise meter project
>> and have acquired a double oven XO that boosts high short term stability
>> (below 1e-12/s). But the spec also states that, even with the double
>> oven, there is still substantial impact of environmental temperature
>> changes (below 1e-8 changes over the normal operating temperature range)
>> so I was wandering if its good practice to try to thermally isolate the
>> DOCXO or do you run the risk of overheating as it always may burn some
>> power and its better to only shield it from draft?
>> 
> 
> I removed an HP 10811A OCXO from a 5370B time interval counter the other
> day and put it into a HP 5352B 40 GHz frequency counter. One thing that
> really struck me is that in the 5370B there was a shroud around the OCXO,
> which is around 5 mm away from the sides of the OCXO. It's made of
> aluminium. But there's nothing like that in the frequency counter. The two
> attached photographs show a significant difference. I took the photograph
> from inside the 5352B frequency counter. The photo of the 5370B was one I
> just found on the EEV blog site, as I did not want to have to mess around
> taking another photograph.
> 
> I see Magnus respond to you.
> 
> My gut feeling is the designers of the 5370B were likely to have more
> knowledge about the behaviour of oscillators than the frequency counter
> designers, which makes me wonder if adding something around the oscillator
> in the frequency counter, like in the 5370B time-interval counter, might be
> a good idea.
> 
> Unfortunately I suspect it would be very time-consuming to evaluate the
> difference a shield would make in the frequency counter, I have another HP
> frequency counter where the fan blows over the oven, which does not seem a
> very good idea.
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[time-nuts] Re: Should a double oven XO be thermally isolated or just draft protected?

[Bob kb8tq via time-nuts]

Hi

Modern OCXO’s are set up based on temperature run data. 
They play with set point (and possibly electrical gain) to optimize
the TC contribution of the crystal *and* the rest of the parts in
the device. This is true of single and double ovens. 

One not so obvious point is that these runs are done in a very
specific temperature environment. Forced air is normally part of 
a test chamber. It also is often part of the OEM installation that 
the OCXO goes into. 

Change the “air pattern” to much and you change the gain of the
oven ( more insulation increases the gain). This can upset the careful
balance done when optimizing the TC of the device. 

One can debate just how stable all of the “stuff” in an OCXO is over
the years. Is a set ( or screening ) done on a production line a decade
ago still relevant today? Random bits of evidence suggest that the TC
optimization holds pretty well, but there isn’t a lot of data. 

A typical double oven should be < 5x10^-10 over 0 to 70C. Indeed 
many manufacturers will sell you examples that are spec’d tighter 
than that. Some offer single ovens with spec’s below 1 ppb over 0 to
70. 1x10^-8 is a very typical single oven spec. 

How well does this or that example do? It is not uncommon to see 
1x10^-8 level single ovens rolling off the production line at <2x10^-9.
On some designs > 80% of the units do this. Counting on any and 
every OCXO to be 5X better than spec …. maybe not, but many designs
do. 

How to “manage” an OCXO? 

First step it to get a good one in the first place. If eBay is your source of 
supply ( it is for me ….) what you get likely is not going to be 100% perfect. 
Some level of testing and sorting will be involved. That needs to be done 
before a lot of additional effort is put in. 

Next up is to plan on keeping it on power all the time. OCXO’s don’t like
to be cycled. Sorry about that. If this bugs you, don’t head down this 
road. There are good reasons for this to bug you so do think about it.

Drafts and abrupt temperature changes are to be avoided. Opening the 
lab window next to your reference standard … not a great idea. Something
as simple as a towel or a cardboard box tossed over the device can do
wonders. Exotic enclosures are probably better, but simple gets you a 
long way. Thermal mass might help as well. 

Just as a note, things like Rb standards (and Masers) also are said to 
benefit from fairly simple “draft protection” enclosures. 

Most folks are pretty obsessive about regulated supplies. If anything they
go a bit overboard in terms of noise for an OCXO supply. What might get
overlooked is the need for a fairly substantial ( = low voltage drop) supply
wiring setup (along with good ground practices). If you plan some sort
of battery backup, consider the regulation impact as it cuts in and out. 

Loading on the output of an OCXO does matter. How much is a “design 
feature”. It is not uncommon to see a few minutes of disruption for a 
significant load change. Simple answer here is not to play with moving 
things around a lot :) 

Many OCXO’s are tuned via an EFC. Feeding this input in a stable fashion
can get a bit crazy. Do try to run the EFC circuit ground straight back to
the OCXO. Oven current induced drops are not great for EFC stability ….

Fun !!!

Bob

> On Jul 1, 2022, at 6:40 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> I'm trying to build a stable reference for a phase noise meter project and 
> have acquired a double oven XO that boosts high short term stability (below 
> 1e-12/s). But the spec also states that, even with the double oven, there is 
> still substantial impact of environmental temperature changes (below 1e-8 
> changes over the normal operating temperature range) so I was wandering if 
> its good practice to try to thermally isolate the DOCXO or do you run the 
> risk of overheating as it always may burn some power and its better to only 
> shield it from draft?
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[time-nuts] Re: What's the best HP OCXO for frequency counter reference?

[Bob kb8tq via time-nuts]

Hi


> On Jun 28, 2022, at 12:15 PM, Hal Murray via time-nuts 
>  wrote:
> 
> 
> Adrian Godwin said:
>> If you use the ovened oscillator for temporary use away from the home GPSDO,
>> how good will the oscillator be with those interruptions to power /
>> temperature, and will it stabilise during the period you're using it there ? 
> 
> You can solve that with a UPS and/or a gizmo that plugs into a car accessory 
> socket.

You still power it down doing that. Power cycle generally isn’t a good idea if 
you 
are after good stability. 

> 
> How much does the mechanical jostling as gear gets moved from bench to car to 
> table effect the frequency?

If you flip it end for end you might be measuring the “2g tip” acceleration 
sensitivity :).
That could be up in the 1 to 2 ppb range. Shock and vibe is very much a “that 
depends”
sort of thing. You could (but probably won’t) see a couple of ppb.

The biggest issue is going to be the changing temperature environment. Thermal 
shock
is going to create a bigger change than simple temperature change. Going from 
inside 
to outside to car to outside to inside …. each one is a relatively fast change. 
You might
have a device that held a ppb in a normal temp test and it changes 10X that (or 
more) 
when hit with a relatively modest “fast change”. 

Just what’s fast and how much of a change you get is very much a “that depends” 
sort
of thing. Leave the cover off your OCXO mechanical trimmer and you could find 
that it
is very sensitive to a gentle gust of wind from the wrong direction ….

Bob

> 
> 
> -- 
> These are my opinions.  I hate spam.
> 
> 
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[time-nuts] Re: What's the best HP OCXO for frequency counter reference?

[Bob kb8tq via time-nuts]

Hi

All HP 10811’s came off the same line and with the exception of the “weird” one 
used
in the original HP GPSDO, they all had the same parts / same optimizations in 
them. The
only thing that was done to make this or that dash number was to screen the 
finished 
units. Once “enough” of this or that spec came out of screening, they stopped 
looking. 
The “next one in line” might have well beat any of the ones in the batch … they 
simply 
had no time ( = funds ) to screen everything. 

Since this screening was done a *long* time ago, there is no way to know if the 
OCXO 
you have still is a “good one”. Stuff happens over time …. subtle changes 
rarely get 
noticed. 

Bob

> On Jun 28, 2022, at 12:03 PM, Dr. David Kirkby 
>  wrote:
> 
> On Tue, 28 Jun 2022 at 20:04, Bob kb8tq via time-nuts 
> mailto:time-nuts@lists.febo.com>> wrote:
> Hi
> 
> The “typical” 10811 struggles when shut down for a while. Once the oven
> is turned off, the boards are just sitting there in whatever environment your
> lab provides. Do they soak up humidity or is it something else? We could 
> (and have) debated that quite a bit. 
> 
> Part of my problem is that if I ever wanted to sell this, I'd like to sell it 
> with an oven that was designed to meet the specifications given in the 
> frequency counter manual for option 010, which are better than the 10811A. 
> 
> Another quite practical issue is setting the blinking frequency, as the pot 
> on the oscillator seems too touchy. I set a signal generator to 20 GHz, which 
> was fed from a GPS frequency reference. The output of the signal generator 
> was then fed into the counter. Getting the counter to read within 100 Hz is 
> extremely difficult, as the tuning control is too coarse. As I write the 
> frequency counter is reading 83 Hz higher than 20 GHz. 
> 
> To be honest, that's "good enough" for what I am practically going to need. 
> But I would like to do better if possible. 
> 
> Dave 

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[time-nuts] Re: What's the best HP OCXO for frequency counter reference?

[Bob kb8tq via time-nuts]

Hi

The “typical” 10811 struggles when shut down for a while. Once the oven
is turned off, the boards are just sitting there in whatever environment your
lab provides. Do they soak up humidity or is it something else? We could 
(and have) debated that quite a bit. 

Regardless of what the issue is, the bottom line is that when you turn the beast
back on again, it *might* take quite a while for it to become reasonably good. 
In some cases that means a couple weeks to hit spec limits. In others it might
mean a week or two to get back to “as good as it was”. 

A *good* 10811 will do much better than what the specs would suggest. It may
be a “one in a hundred” sort of item, but they do exist. Ones that barely / 
don’t 
meet spec (even after a month on power) also exist and they are a lot more 
common 
than the ones we brag about with various plots. 

What’s the point? Keeping the unit on power is a good idea, even if you have
a good example. Keeping a poor one on power still will not turn it into a good 
one. Checking out the units you have is a “needed thing” before you start 
counting
on them for this or that. They *all* left the factory a *long* while ago. 

It *does* matter just what you are counting on the OCXO to do. Does phase 
noise at 100 KHz offset matter in your case? Does 10 second ADEV at 5x10^-12 
degrade your performance compared to 5x10^-13? Is drift the only thing that 
really 
matters? ( and when did you last set it on? :) :) ) If so, temperature 
stability (in a 
“transport someplace” situation)  likely also matters, testing it is not 
something 
most are set up to properly do ….

One alternative: Grab a fairly typical telecom Rb. Put a (cheap) cleanup 
oscillator 
on it. Use that as your “portable” reference. Run it on batteries (or a 12V 
plug in 
the car)  as you transport this or that item here or there. Let it sit 
unpowered on 
the shelf until about a week ahead of your planned expedition. 

Fun !!!

Bob

> On Jun 28, 2022, at 9:20 AM, Adrian Godwin via time-nuts 
>  wrote:
> 
> Related to that ..
> 
> If you use the ovened oscillator for temporary use away from the home
> GPSDO, how good will the oscillator be with those interruptions to power /
> temperature, and will it stabilise during the period you're using it there ?
> 
> I don't know what the vales are, but I'd suggest the option 010 will never
> reach the 001's spec and the 001 may not even be justified.
> 
> 
> 
> On Tue, Jun 28, 2022 at 6:13 PM Dr. David Kirkby via time-nuts <
> time-nuts@lists.febo.com> wrote:
> 
>> I have a 5352B 40 GHz frequency counter which was fitted with a TCXO. I
>> removed that and fitted an HP 10811-60111 S/N 2332A17049 which I removed
>> from a 5370B time interval counter - I have a few of those, and the
>> microwave frequency counter needed the oven more than the time-interval
>> counter.
>> 
>> Looking at the specification of the 5352B, there were 3 oscillator options
>> 
>> * Standard TCXO
>> * Option 001 Oven time based. Long-term aging < 5 x 10^-10 / day after 24
>> hour warmup. < 1 x 10^-7 / year for continuous operation.
>> * Option 010 High stability time base. Long term aging < 1 x 10^-10 / day.
>> < 3.6 x 10^-8 / year for continuous operation.
>> 
>> To be honest, when the instrument is here, I will use a GPS reference. But
>> I might want to take it to the amateur radio club sometimes. I would like
>> to get the best oscillator I can. Are any models going to be better than
>> others, or by this time, is it just pot luck? I think it's the latter, but
>> maybe some are double-ovens and some single.
>> 
>> With the exception of power cuts of up to a few hours, the HP 10811-60111 I
>> fitted has been continuously powered on for a few years. But due to soaring
>> power costs, I am going to switch my ovens off.
>> 
>> Dr David Kirkby Ph.D
>> Email: drkir...@kirkbymicrowave.co.uk Web:
>> https://www.kirkbymicrowave.co.uk/
>> Kirkby Microwave Ltd (Tel 01621-680100 / +44 1621-680100)
>> Stokes Hall Lodge, Burnham Rd, Chelmsford, Essex, CM3 6DT.
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[time-nuts] Re: First PN measurement results at 1 Hz to 20 kHz from carrier

[Bob kb8tq via time-nuts]

Hi

> On Jun 27, 2022, at 1:43 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Magnus, Bob,
> When the mixer is operating in the linear region for the DUT input (0dBm or 
> lower), would it be possible to use a calibrated noise sources  to do an 
> extra verification of the noise level measurement?
> Of course with a noise source you get 3dB as both sidebands fold.

The “normal” approach is to put the mixer into saturation. This gives you the
best noise floor. It also does a bit better at separating AM noise from PM noise
(since you are trying to measure phase noise …..).

> Verification steps:
> Verify the DUT output level is correctly brought to 0dB (using attenuators) 
> using a calibrated spectrum analyzer
> Connect the DUT to the phase measurement setup and set the reference to a 
> 500Hz offset to get a beat note and verify the beat note is registered at 
> 0dB, change the DUT level some dB up and down to confirm its in a linear 
> region.
> Measure the per Hz output power of a noise source using a calibrated spectrum 
> analyzer and a noise marker set to 10MHz.
> Connect the noise source to the phase measurement setup and check if the 
> noise level is measured at level measured by the spectrum analyzer + 3dB
> This should work if the RBW of the phase measurement is indeed set to 1Hz.

If you do it this way, you still need to do conversion for the “one radian” 
reference
level that is used with phase modulation ( = the reference is *not* one cycle 
). Yes
that’s a bit weird / obscure.


> 
> Another verification option may be to use the phase modulation of a signal 
> generator. This can not check the effective noise bandwidth of the FFT but it 
> can check linearity over the whole range.
> The output of the mixer is terminated with 50ohm so a factor of 10 in voltage 
> should give a 20dB power step.

Audio termination at 50 ohms does not do much for isolation. (again a bit of an 
obscure
point). By terminating in 10X the nominal impedance ( so 500 ohms in this case) 
you get
another 6 db of gain in the system. Since this is ahead of the preamp, it might 
improve 
your noise floor. 

Bob


> When operation in the linear range the phase noise measurement setup should 
> measure 20dB less with every factor 10 reduction in phase modulation depth 
> where 90 degrees is equal to 100% modulation depth so equal to the signal you 
> get when measuring a beat note.
> When measuring with modulation depth of 90,9,0.9,0.09 and 0.009 degrees the 
> measured level should step from 0,-20,-40, -60 to -80dB
> 
> Any feedback?
> Erik.
> 
> On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
>> Hi Erik!
>> 
>> Great progress! Sure interesting to look at them phase-noise plots, right? 
>> It's a really good tool in addition to the stability of ADEV and friends.
>> 
>> As I recall it, the ADE-1 is not documented to be isolated, but it is very 
>> obvious when you look down the backside of it. However, it has capacitive 
>> coupling and one should consider both common mode rejection and common mode 
>> loading it down for these to work well.
>> 
>> Word of caution when it comes to levels, as the windowing filter used causes 
>> shifts in noise-levels, so estimation of noise-levels becomes a little bit 
>> tricky as you try to get the nitty gritty right, but getting the overall 
>> shape view you already gained a lot with the things you achieved.
>> 
>> A technique used to push further down into lower noise-levels is the 
>> cross-correlation technique, where you split the signal into two channels, 
>> each being exactly what you have now, and then rather than squaring the 
>> output of the FFT from each channel, you multiply one with the completment 
>> of the other, then average on those. This allows you to supress the noise of 
>> each reference oscillator. You do not have to go there from start, as you 
>> already make very useful measurements, but I'm just suggesting what may lie 
>> up ahead.
>> 
>> Compared to some of the other sources, the Rigol SG does fairly well, but 
>> then again, things can be even more quiet. For the XO you can see the 15 
>> dB/Oct slope as expected for flicker frequency. Try to locate the source of 
>> the peaks you see and see if you can clean it up. The XO seems to be a 
>> fairly good DUT for doing that.
>> 
>> Cheers,
>> Magnus
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[time-nuts] Re: First PN measurement results at 1 Hz to 20 kHz from carrier

[Bob kb8tq via time-nuts]

Hi

Ok, so mixer phase noise calibration:

Set things up with “full blast” inputs to both sides of the mixer. Keep them
at the same level through all tests. This might be +7 on both ports, it might
be +13, it could be +20 dbm. ( yes, mixers *do* get fried this way ….). 

First cal step, you have the radians to volts factor. The “zero db” point for 
phase 
noise is one radian. It is unlikely that you come up with exactly one volt per
radian. Oddly enough …. it can happen. This is done by looking at a free running
beat note and measuring the slope at a zero crossing. ( 0.1 volts over 10 
degrees
maybe …). If you have a pre-amp involved, it needs to be fiddled to keep it out 
of saturation. 

Back in the day of analog devices, you had a bunch of fun with bandwidths
and averaging factors. These days you run a program and it gives you a 1 hertz
normalized number. You are missing a *lot* of fun there …. :)

As previously mentioned, your result is at least 3 db higher than the single 
sideband phase noise. ( Yes, that’s the definition … single sideband). Back 
before the modern era folks used 6 db close in. Not so much anymore.

Finally, you get whatever the combination factor is for your two sources. If 
they are identical, you get another 3 db. 

Spurs are a bit of an issue. Since you have done this or that for a 1 Hz 
normalization, that can mess up the spur levels. They are an absolute number 
rather than bandwidth normalized. For home / basement use, you typically ignore 
this. On a “pro” device, they have fancy software that guesses what is a spur 
and 
de-normalizes that section.

Some folks do the whole slope measurement thing every time. Others do a 
(unsaturated preamp) beat note and then use a “known” correction factor. Since 
the beat note is not a sine wave, it can not be used directly as a reference. 
The “right” 
answer is to do the slope every time. If you are measuring a hundred same / 
same 
devices that day, you go with the beat note. 

Bob

> On Jun 25, 2022, at 10:07 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Thanks to all the great help from people on this list I was able to make some 
> progress in doing close-in phase noise measurements.
> The setup consists of a VC-OCXO going into the LO port of  an ADE-1 mixer, 
> The DUT into the RF port, the IF port is low pass filtered and used to steer 
> the VC-OCXO and is send to a high quality 24 bit USB audio capture unit 
> connected to a PC running ARTA.
> The ADE-1 mixer was selected because that all ports are completely isolated 
> from each other, there is no common ground, which helps to reduce ground loop 
> problems a bit.
> The log plots from ARTA confirm a 130dB dynamic range and the resolution 
> bandwidth is supposed to be about 1Hz. Each plot was averaged over 10 
> measurements
> All input signals where normalized by attenuators to have their carrier at 
> 0dB in the plot.
> Using a generator at variable frequency offset it was confirmed the audio 
> input is flat down to 1Hz.
> Using a generator with phase modulation down to 0.001 degree the sensitivity 
> of the measurement chain was checked. (20dB level reduction with every factor 
> 10 reduction in phase modulation depth)
> It is expected to have at least +/-5dB level inaccuracy.
> The DUTS measured where:
> - a fairly clean XO (PN_XO.JPG)
> - a rather bad GPSDO output (PN_GPSDO.JPG)
> - The not so famous cheap Chinese TCXO (PN_TCXO.KPG)
> - The output of a Rigol SG (PN_Rigol.JPG)
> The XO is the cleanest
> The TCXO shows odd spurs between 10 and 40 Hz and the PN does not drop down 
> as it should (spec states: -135dBc/Hz at 1kHz offset)
> The GPSDO is terrible, this demonstrates you can have a 1e-10 ADEV at 1s tau 
> from a bad oscillator.
> The Rigol is not so clean and a PLL shoulder seems to be present just above 
> 1kHz.
> 
> Next step is to add low noise gain close to the mixer LPF output to get more 
> dynamic range and a better VC-OCXO (Morion MV170 (PN -100dBc/Hz at 1Hz 
> offset) to lower the impact of the reference VC-OCXO
> Erik.
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[time-nuts] Re: Knowing when the PPS signal is locked to the satellite.

[Bob kb8tq via time-nuts]

Hi

Keep in mind that if you run the PPS when unlocked, the uBlox does not 
“play nice” when it locks up. The PPS moves to where ever it needs to be
quite abruptly. That may or may not create issues, depending on your 
application.

Bob

> On Jun 22, 2022, at 4:59 AM, Dave via time-nuts  
> wrote:
> 
> I am using a NEO-6M to provide a PPS signal for an experiment but I need to 
> know when it is locked or not.
> 
> Does anyone know if there is a way to find out from either the NMEA data 
> stream or via the proprietary protocol ??
> 
> At the moment I have it programmed to only provide a PPS signal when it is 
> locked (or nothing if not) and monitor
> the PPS signal in the microprocessor, but it's not entirely satisfactory and 
> I need a better mechanism.
> 
> Dave
> 
> 
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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

With an audio spectrum analyzer, and an RPD-1 mixer, and +10 dbm on 
each side of the mixer …. you should be able to get away with about 20 db
of “preamp gain”. Yes, that’s dependent on exactly what analyzer you have.

The typical sound card may well need closer to 50 db of gain. That’s better 
done with two op-amp stages than trying to get it all in one shot. Again there’s
an assumption involved about wanting 40 to 100 KHz sort of “top end” for the
measurement. 

Ground loops ( and power line noise ) are indeed a *very* big thing in all these
measurements. ADEV / phase noise / whatever. Being able to see the crud makes
getting rid of it much easier. That’s not to in any way to say it’s easy, only 
that it’s
easier. 

Fun !!!

Bob

> On Jun 21, 2022, at 1:26 AM, Erik Kaashoek  wrote:
> 
> Bob, Using the approach you described I was able to verify the noise floor of 
> my initial measurement setup to be at -80dBc at 10Hz offset from carrier and 
> -100dBc above 1kHz with a strong peaks at 50Hz, 60Hz and harmonics. This all 
> with an RBW in the FFT of around 4Hz.
> This level of noise is way to high to do phase noise measurement so I'm now 
> going to work on removing ground loops and adding low noise amplification.
> Erik.
> 
> On 21-6-2022 2:04, Bob kb8tq wrote:
>> Hi
>> 
>> Ok, single mixer phase noise basics:
>> 
>> First thing is to womp the mixer up to the point it almost smokes. Putting 
>> +7 dbm into
>> both ports on a “7 dbm” mixer is very normal in this case. Watching for the 
>> fact that the
>> mixer likely is *not* a 50 ohm load is part of the process ( = pads might 
>> help out) as well
>> as understanding that it does not have a monster amount of isolation ( = 
>> isolation amps
>> may be needed ).
>> 
>> Next one generates a beat note by offsetting the two oscillators a bit. This 
>> gives you a nice
>> 360 degree sweep function ( 360 degrees per cycle :) ). From that you can 
>> work out the
>> system sensitivity in volts per degree ( or better yet per radian since 
>> that’s what you actually
>> want as the “magic number” …. it’s phase modulation so radian is king …).
>> 
>> Next you lock the two oscillators together via a DC feed out of the mixer to 
>> one or the
>> other of them. You adjust the “lock point” so that it is at zero volts out 
>> of the mixer. This
>> puts the two oscillators in quadrature. Yes, there is that messy 2X the 
>> input frequency RF
>> output and the inevitable leakage. Those are handled with a lowpass filter.
>> 
>> The output of the mixer is now “just noise”. There is no nasty carrier to 
>> deal with. There is
>> no messy fold over to wonder about. What you get is the DSB noise ( so both 
>> sides of
>> carrier) from the sum of the two oscillators.
>> 
>> Output of the mixer goes up if you terminate it in an “high” load. Something 
>> like 500 ohms
>> on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems 
>> to be ok
>> either way and the added gain / better floor is “free”.
>> 
>> Simply put you add 3 db when you look at DSB if it’s uncorrelated, and 
>> another 3 db if the
>> oscillators are identical. Your “output” is 6 db higher than the single 
>> sideband / single oscillator
>> phase noise. You can argue that close in noise is likely correlated due to 
>> it being a modulation
>> on the carrier. The standard convention is to use 3 db.
>> 
>> Amplify the noise up and you can measure very low levels of phase noise. Low 
>> noise
>> audio op-amps are pretty easy to find spec sheets on. With anything these 
>> days finding
>> them on the shelf may be “interesting”. The OP-27 / OP-37 with low 
>> resistance in the
>> feedback path go way back for this application. There are a lot of other 
>> candidates.
>> 
>> The cutoff of the lock signal typically is adjustable to keep it below the 
>> lowest point
>> of interest for your noise testing. If that is impractical, there are ways 
>> to calibrate and
>> read “inside the loop”.
>> 
>> The HP 3048 phase noise analyzer was based on this approach. The original 
>> app note most
>> folks started from came from Fluke back in the early 1970’s. I have not 
>> (yet) found a good
>> copy of it on the internet.
>> 
>> Fun !!!
>> 
>> Bob
>> 
>> 
>> 
>>> On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts 
>>>  wrote:
>>> 
>>> Bob, Magnus,
>>> Thanks, clear. A counter is for ADEV, not for phase noise.
>>> I made a test setup to learn how 

[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

Ok, single mixer phase noise basics:

First thing is to womp the mixer up to the point it almost smokes. Putting +7 
dbm into
both ports on a “7 dbm” mixer is very normal in this case. Watching for the 
fact that the
mixer likely is *not* a 50 ohm load is part of the process ( = pads might help 
out) as well 
as understanding that it does not have a monster amount of isolation ( = 
isolation amps 
may be needed ).

Next one generates a beat note by offsetting the two oscillators a bit. This 
gives you a nice 
360 degree sweep function ( 360 degrees per cycle :) ). From that you can work 
out the 
system sensitivity in volts per degree ( or better yet per radian since that’s 
what you actually
want as the “magic number” …. it’s phase modulation so radian is king …).

Next you lock the two oscillators together via a DC feed out of the mixer to 
one or the 
other of them. You adjust the “lock point” so that it is at zero volts out of 
the mixer. This
puts the two oscillators in quadrature. Yes, there is that messy 2X the input 
frequency RF
output and the inevitable leakage. Those are handled with a lowpass filter. 

The output of the mixer is now “just noise”. There is no nasty carrier to deal 
with. There is 
no messy fold over to wonder about. What you get is the DSB noise ( so both 
sides of
carrier) from the sum of the two oscillators. 

Output of the mixer goes up if you terminate it in an “high” load. Something 
like 500 ohms
on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to 
be ok
either way and the added gain / better floor is “free”. 

Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 
3 db if the 
oscillators are identical. Your “output” is 6 db higher than the single 
sideband / single oscillator 
phase noise. You can argue that close in noise is likely correlated due to it 
being a modulation
on the carrier. The standard convention is to use 3 db.

Amplify the noise up and you can measure very low levels of phase noise. Low 
noise
audio op-amps are pretty easy to find spec sheets on. With anything these days 
finding
them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance 
in the 
feedback path go way back for this application. There are a lot of other 
candidates. 

The cutoff of the lock signal typically is adjustable to keep it below the 
lowest point 
of interest for your noise testing. If that is impractical, there are ways to 
calibrate and 
read “inside the loop”. 

The HP 3048 phase noise analyzer was based on this approach. The original app 
note most
folks started from came from Fluke back in the early 1970’s. I have not (yet) 
found a good
copy of it on the internet. 

Fun !!!

Bob



> On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Bob, Magnus,
> Thanks, clear. A counter is for ADEV, not for phase noise.
> I made a test setup to learn how to use the mixer/PLL approach.
> First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC 
> shift with shifting the phase between the two signal.
> Then by modulating one output with FM or PM.
> There is a low pass filter after the mixer to get rid of the 10 MHz and its 
> harmonics but the LPF is measured flat till about 10kHz.
> The output signal from the mixer was kept within 10% of the full voltage 
> swing to stay in the (hopefully) linear range.
> Using PM creates a low frequency output from the mixer that is proportional 
> to the phase shift (region 0-1 degree) and constant in amplitude with change 
> of frequency. Also when using external modulation from an audio signal 
> generator created the expected behavior with drive level and no frequency 
> impact
> Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases 
> very fast with increasing frequency (range 0.1 to 10 Hz)
> Also when using 1 Hz or more frequency deviation. The higher frequency 
> deviation leads to higher output levels as expected.
> Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation 
> and 0.1 to 1 Hz frequency deviation) translates to the calibration example 
> mentioned in the document on phase noise measurement as linked by Bob. (0.1 
> Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
> At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a 
> lot of output. Why is this output frequency dependency?
> Is this a problem with the signal generator?  Or the mixer?
> Then I tried to use the modulated signal from the SG PLL locked to a 10MHz 
> VCO. Results where the same. FM output signal is frequency dependent, PM not.
> Erik.
> 
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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

A “proper” phase noise analyzer will get you down to < -165 dbc/Hz. It also 
will preserve
the frequency spectrum ( no sampling / nyquist roll over ). What you get at 132 
Hz offset 
is the (DSB) noise at that offset and *only* that noise. 

With a counter ( as Magnus mentions ) the sampling process looks at and sums up 
all the
noise in the input bandwidth. If it’s a 0.01 second sample, you get noise from 
100Hz, 200Hz,
300Hz on out to whatever the counter’s input bandwidth is. 

Since the counter likely has a bandwidth of a couple hundred MHz and you are 
measuring
something like 10 or 25 MHz, you sum up a whole lot of “noise floor” noise. 
Even if the counter
input amp has a zero db noise figure and the sampler does as well, you “fold 
in” 200 MHz worth
of that noise. This makes the sensitivity of the counter to low levels of phase 
noise pretty poor. 

If you didn’t have this sort of limitation, your counter might well have 20 fs 
resolution rather than 
20 ps …. (and no, this isn’t the only reason you have limited resolution)

Bob

> On Jun 19, 2022, at 10:45 PM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Bob,
> Many thanks for the guidance you provide and the phase noise measurement 
> document.
> Can you provide feedback on this reasoning: A counter is like an ADC but in 
> the frequency domain. So if you measure with 0.01 s tau you basically average 
> over 0.01 s so you can only observe "phase noise" (e.g. energy that is not at 
> the exact requested frequency) up to maximum 50 Hz from the carrier. But as 
> you measure the true frequency changes the sensitivity of this measurement is 
> extremely high. Translating the amount of time spend at a certain frequency 
> away from the carrier (ADEV?) into a phase noise number in dBc is something I 
> do not yet understand.
> With a (very good) spectrum analyzer you may be able to come close to the 
> carrier but as there is so much energy in the carrier it will be difficult to 
> observe phase noise energy closer than say 1 or 10 kHz (at least not with the 
> equipment I can afford) so any phase noise plot created using a spectrum 
> analyzer can not be better than the combined phase noise of all LO's in the 
> spectrum analyzer and will start at say 1 or 10 kHz.
> For the frequencies between 50 Hz and 20 kHz the simplest option is to use a 
> second LO and a mixer and a slow  (loop BW below 10 Hz)PLL to keep the mixer 
> in quadrature and feed the output of the mixer, after low pass filtering, 
> into a PC soundcard for FFT processing.
> Erik.
> 
> On 19-6-2022 22:45, Bob kb8tq via time-nuts wrote:
>> Hi
>> 
>> As HP found out back around 1973 or so, translating ADEV to phase noise
>> is not possible. This is true, even if you have the ADEV numbers for a 
>> variety
>> of Tau values as opposed to some sort of “average” kind of number.
>> 
>> There are a number of things ( like spurs ) that can strongly influence a 
>> counter
>> based ADEV reading, and have very little impact on a phase noise ( or signal 
>> to
>> noise reading.  There also are ways the shape of the phase noise curve can
>> impact ADEV and have very little signal to noise impact for a specific 
>> signal.
>> 
>> By far the best way to do this is to properly measure phase noise at various
>> offsets from carrier. You can then look at the dbc/Hz numbers at each offset.
>> This lets you see what your devices are doing to the signal. You can then 
>> track
>> down the offending bit or piece and fix the problem.
>> 
>> The easiest way I know of to do phase noise is to quadrature lock two 
>> identical
>> sources into a double balanced mixer. You then put in a simple amplifier 
>> stage
>> to drive the mix down output into a sound card or spectrum analyzer. Total 
>> cost
>> if you already have a sound card should be < $50 ( US dollars …) for a DIY 
>> version.
>> That assumes you have the usual junk box parts and do a point to point wire
>> version.
>> 
>> Some example ADEV plots:
>> 
>> http://leapsecond.com/museum/manyadev.gif 
>> <http://leapsecond.com/museum/manyadev.gif>
>> 
>> http://leapsecond.com/museum/manyadev.gif 
>> <http://leapsecond.com/museum/manyadev.gif>
>> 
>> Some plots of a number of measurements:
>> 
>> http://www.leapsecond.com/pages/fe405/ 
>> <http://www.leapsecond.com/pages/fe405/>
>> 
>> Quick primer on phase noise measurement
>> 
>> https://www.npl.co.uk/special-pages/guides/gpg68_noise 
>> <https://www.npl.co.uk/special-pages/guides/gpg68_noise>
>> 
>> ( The easy approach starts on page 21 :) )
>> 
>> Bob
>> 
>> 
&

[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

As HP found out back around 1973 or so, translating ADEV to phase noise 
is not possible. This is true, even if you have the ADEV numbers for a variety
of Tau values as opposed to some sort of “average” kind of number.

There are a number of things ( like spurs ) that can strongly influence a 
counter
based ADEV reading, and have very little impact on a phase noise ( or signal to
noise reading.  There also are ways the shape of the phase noise curve can
impact ADEV and have very little signal to noise impact for a specific signal. 

By far the best way to do this is to properly measure phase noise at various 
offsets from carrier. You can then look at the dbc/Hz numbers at each offset. 
This lets you see what your devices are doing to the signal. You can then track
down the offending bit or piece and fix the problem. 

The easiest way I know of to do phase noise is to quadrature lock two identical
sources into a double balanced mixer. You then put in a simple amplifier stage
to drive the mix down output into a sound card or spectrum analyzer. Total cost
if you already have a sound card should be < $50 ( US dollars …) for a DIY 
version.
That assumes you have the usual junk box parts and do a point to point wire
version. 

Some example ADEV plots:

http://leapsecond.com/museum/manyadev.gif 


http://leapsecond.com/museum/manyadev.gif 


Some plots of a number of measurements:

http://www.leapsecond.com/pages/fe405/ 

Quick primer on phase noise measurement 

https://www.npl.co.uk/special-pages/guides/gpg68_noise 


( The easy approach starts on page 21 :) )

Bob


> On Jun 19, 2022, at 11:40 AM, Karen Tadevosyan via time-nuts 
>  wrote:
> 
> Hi 
> 
> 
> 
> Thank you for the clarification and rf-tools link.
> 
> 
> 
> Agree with your calculation. That’s why I raised this question regarding a 
> fixing PN degradation by Pendulum CNT-91.
> 
> 
> 
> Could you please explain where is the error in my reasoning of the experiment 
> :
> 
> 
> 
> * There is one 10 MHz OCXO with ADEV = 5 mHz
> * There are two boards (DUT1 and DUT2) which multiply 10 MHz OCXO signal 
> by 2.5 using the PLL method
> * DUT1 has 25 MHz output signal with high PN  (checking by air and by 
> measurement of S/N)
> * DUT2 has 25 MHz  output signal with low PN  (checking by air and by 
> measurement of S/N)
> Experiment’s steps:
> * Step 1: DUT1 ADEV measuring gives me a value of 60 - 70 mHz instead of 
> the expected 12.5 mHz  (5 mHz x 2.5)
> * Step 2: DUT2 ADEV measuring gives me a value of 10 - 12 mHz which 
> matches the expected 12.5 mHz  (5 mHz x 2.5)
> * Step 3: based on ADEV values which in the first case (DUT1) are much 
> greater than expected and in the second case (DUT2) coincide with the 
> expected I conclude that PN of the output signal from DUT2 will be lower than 
> from DUT1.
> I can see this PN degradation using Pendulum CNT-91 only as R FSQ8 does not 
> fixate any PN degradation between DUT1 and DUT2
> 
> Karen, ra3apw
> 
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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

If the objective is “phase noise -130 dbc at a 10 KHz offset” and the signal is 
at 25 MHz, the resultant jitter ( in a 10 KHz bandwidth ) is likely in the < 
500 fs 
range. The CNT-91 is only a 50 ps resolution device. 

Since it’s a counter, there is no practical way to vary the bandwidth. You get 
whatever the internal processing gives you. Even then, at 25 MHz a broadband
-130 dbc/Hz phase noise ( 10 MHz bandwidth) is still below 10 ps. Still below 
the counter’s floor. 

One of *many* calculators that let you play with jitter from phase noise:

https://rf-tools.com/jitter/ 

It’s just the first one Mr Google turned up. There may be much better ones
out there. 

Bob

> On Jun 19, 2022, at 9:58 AM, Karen Tadevosyan  wrote:
> 
> Hi
> 
>> The CNT-91 has a very high noise floor if you are trying to “see” phase 
>> noise.
> 
> May be but with this Pendulum CNT-91 I can fix PN degradation in the range 
> -130 ... -150 dBc/Hz where R FSQ8 can do nothing...
> 
> FYI: FSQ8 is not the worst SA: PN floor -130 dBc/Hz @10 kHz on 1 GHz.
> 
> Karen, ra3apw   
> 
> 
> 

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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

The CNT-91 has a very high noise floor if you are trying to “see”
phase noise. It also does not give you much in the way of frequency
offset information. A single mixer setup running into a sound card
is a *much* better approach if you are looking for another opinion 
vs the FSQ8. I have used the CNT 90 and CNT 91’s a lot. I have no
experience with the FSQ8.

Bob

> On Jun 18, 2022, at 10:42 PM, Karen Tadevosyan  wrote:
> 
> Hi Bob,
> 
> Thank you for the advice. An issue with PN of  25 MHz reference source has 
> been decided - I found the PN source, changed schematic and now all works 
> well. 
> 
> My question is about an opportunity to fix PN degradation by Pendulum CNT-91 
> frequency meter-analyzer (which I measure the Allan deviation) which allow to 
> record significantly lower PN values than R FSQ8.
> 
> Karen, ra3apw
> 
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[time-nuts] Re: Fixing PN degradation via ADEV measurement

[Bob kb8tq via time-nuts]

Hi

The “typical” approach to doing this is to multiply up to something around
100 MHz with a relatively narrowband PLL. You can get ( or build ) VCXO’s
in this range with phase noise (at 10KHz offset) in the 160’s to (maybe) 170’s. 
This is much better performance than a 5 or 10 MHz reference can deliver
when multiplied to that frequency. 

The “cross over” between multiplied reference noise and a fairly good VCXO
should typically be in the 50 to 200 Hz range. Since it is highly dependent on 
the exact parts chosen, there is no “one size fits all” answer.

Bob

> On Jun 18, 2022, at 12:49 AM, Karen Tadevosyan via time-nuts 
>  wrote:
> 
> Hello group,
> 
> 
> 
> I am developing a hamradio ground satellite station for the first 
> geostationary satellite QO-100 (uplink 2.4 GHz / downlink 10 GHz).
> 
> 
> 
> Narrowband digital communication requires high frequency stability so I use a 
> 10 MHz OCXO with an absolute Allan deviation (ADEV) of about 5 mHz as a 
> single reference oscillator.
> 
> 
> 
> To form a reference signal of 25 MHz for the 9750 MHz synthesizer in LNB I 
> use a PLL (to multiply 10 MHz OCXO frequency by 2.5 times).
> 
> 
> 
> When measuring the PN of a shaped reference signal at 25 MHz, measured with 
> SA R FSQ8 achieved an instrument limit of -130 dBc/Hz @ 10 kHz. According 
> to my calculations, the 25 MHz output PN should be in the range of -150 
> dBc/Hz. However, when operating via satellite, the nature of the received 
> signal is more "noisy and smeared" with a lower S/N ratio (compared to 
> operation from another low-noise generator at 25 MHz without PLL) which, in 
> my opinion, indicates a worse PN value than expected -150 dBc/Hz.
> 
> 
> 
> At the same time I noticed such a feature - when measuring the absolute ADEV 
> value of 25 MHz output signal the device does not show the calculated value 
> of 5 MHz x 2.5 = 12.5 mHz but a value of approximately 60 - 70 mHz.
> 
> 
> 
> Additional tests at 10 GHz through the radio end-to-end confirmed a real 
> decrease in S/N ratio of about 10-15 dB and I found the source of the problem.
> 
> 
> 
> Question: it turns out that the Pendulum CNT-91 frequency meter-analyzer 
> (which I measure the Allan deviation) allow to record significantly lower PN 
> values ​​than the R FSQ8?
> 
> 
> 
> That is, with PN borderline values between -130 and - 150 dBc / Hz (in 
> our particular case), the Allan deviation measurement allow to accurately fix 
> the PN degradation where a not the worst SA R FSQ8 can not help.
> 
> 
> 
> I would like to hear the opinion of experts in this matter.
> 
> 
> 
> Karen, ra3apw
> 
> 
> 
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

Like it or not, modern crystal packages are mad of metal ( as opposed to glass).
The gotcha with metal is that getting rid of *all* of the He is darn hard. That 
makes
a “perfect vacuum” more than a bit tough.

Bob

> On Jun 12, 2022, at 7:38 PM, Lux, Jim via time-nuts 
>  wrote:
> 
> On 6/12/22 6:30 PM, Bob kb8tq via time-nuts wrote:
>> Hi
>> 
>> Tear into some of your SC cut based OCXO’s. Take a look at the crystal 
>> package. For
>> bonus points, open up the crystal package. If you have the gear to test it, 
>> take a look
>> at what the gas *is* inside the package. ( Good luck with that :) :) :) )
>> 
>> If you had the gear and the willingness to scrap out OCXO’s you would find 
>> that a number
>> of fast warmup OCXO’s have a *tiny* amount of He in the package. Measuring 
>> this would
>> be tough ( it’s that small). Go through the thermal modeling and it’s *way* 
>> more conductive
>> (thermal wise) than a *perfect* vacuum ……
>> 
>> Bob
> 
> 
> And the worst thing is that if your vacuum sealed widget is in an atmosphere 
> with more He around (like waiting for a launch, in a place that does more He 
> leak tests, etc.), the He will diffuse into your package and it doesn't work 
> like expected.
> 
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

Tear into some of your SC cut based OCXO’s. Take a look at the crystal package. 
For 
bonus points, open up the crystal package. If you have the gear to test it, 
take a look 
at what the gas *is* inside the package. ( Good luck with that :) :) :) )

If you had the gear and the willingness to scrap out OCXO’s you would find that 
a number
of fast warmup OCXO’s have a *tiny* amount of He in the package. Measuring this 
would 
be tough ( it’s that small). Go through the thermal modeling and it’s *way* 
more conductive 
(thermal wise) than a *perfect* vacuum ……

Bob

> On Jun 12, 2022, at 9:18 AM, Ross P via time-nuts  
> wrote:
> 
> I have seen that manufacturers seal their crystals in a vacuum, maybe air 
> interaction affects Q. The point that vacuum inhibits heat flow is something 
> I have never considered in ovenized units. My ovenized crystals take about an 
> hour to settle. I have some WW2 surplus crystals in non-sealed packages that 
> I have not tested... something to do.rp
> 
>On Sunday, June 12, 2022 at 07:26:19 AM PDT, Louis Taber via time-nuts 
>  wrote:  
> 
> I have been of the impression for years now that most "better" crystals are
> in a vacuum.  And the electrical and mechanical connections to the quartz
> itself place as little mechanical load on the crystal as possible.
> Thermal conductivity from the oven to the crystal itself would be both
> hard to model and hard to speed up.
> 
> IR transmission of energy to the crystal also seems problematic considering
> the IR transmission of quartz and the IR reflectivity of gold
> contact plating.
> 
> Is any of this an issue?
> 
>   - Louis
> 
> On Fri, Jun 10, 2022 at 9:53 PM Bob kb8tq via time-nuts <
> time-nuts@lists.febo.com> wrote:
> 
>> Hi
>> 
>>> On Jun 10, 2022, at 2:38 PM, Lux, Jim via time-nuts <
>> time-nuts@lists.febo.com> wrote:
>>> 
>>> On 6/10/22 1:57 PM, Dr. David Kirkby wrote:
>>>> On Fri, 10 Jun 2022 at 17:39, Lux, Jim via time-nuts <
>> time-nuts@lists.febo.com> wrote:
>>>> 
>>>> On the subject of rapid warm up. I suppose if you had a need, one
>>>> could
>>>> dump as much power as you need into the heater. Turn on oscillator,
>>>> lights in room dim for a few moments.
>>>> 
>>>> 
>>>> Is that not likely to damage a crystal? Different parts of the crystal
>> and likely to be at significantly different temperatures at the same time,
>> putting a lot of stress on the crystal due to a thermal gradient. It's
>> probably a bit academic, as nobody is going to make an oven that heats up
>> in fractions of a second, but if one did, I suspect it might not do the
>> crystal a lot of good. This is only an educated guess - I don't have
>> anything to back it up.
>>> Oh, it would be disastrous, although quartz is pretty strong, all the
>> rest of the mounting components might not be.
>> 
>> Indeed, breaking a quartz blank via thermal stress would be very hard to
>> do.
>> The “rest of the parts” actually are pretty durable as well. Most of it is
>> metal and
>> it is quite able to handle thermal issues.
>> 
>> The big issue in a fast warm up AT turned out to be designing the heater
>> and the
>> mount to get the energy to the blank quickly….. If you use a small enough
>> package
>> and blank, the amount of power turns out to be surprisingly small.
>> 
>> If you want to go bonkers, you mount the heaters *inside* the crystal
>> package. This
>> does indeed create some issues in various areas.
>> 
>> Bob
>> 
>>>> 
>>>> At the other extreme,  would there be any advantage in actually heating
>> the crystal very slowly, over the course of an hour/day/week, so the
>> temperature gradient across the crystal is very small? Of course, if an
>> oven took ages to reach the correct temperature, it would be inconvenient
>> for most applications, but for some applications, the advantages might
>> outweigh the disadvantages. Of course, if one does this, I suspect one
>> would have to cool the crystal slowly too to prevent a significant thermal
>> gradient across the crystal.
>>>> 
>>>> I know it's a bit different, but I have a 600 mm f4 Nikon camera lens.
>> I was told that Nikon cools the front element over a period of 6 months to
>> reduce stresses in the glass.
>>> 
>>> Big glass mirrors for telescopes do the same.
>>> 
>>> 
>>> 
>>>> 
>>>> Dave
>>> 
>>> ___
&g

[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

Well ….. 

The question becomes “how much of a vacuum? I surprisingly small amount
of He will have a substantial impact on heat flow. Since there will always be 
some
leakage / outgassing in any package design, it just *might* improve aging.

The days of “spring wire” mounts are long gone for precision crystals. The 
mounts
are short fat metal “bands” that will move heat pretty well. On the typical 
“TO” style
package (HC-40 etc…) the thermal path isn’t all that bad. 

The location of the heaters *is* under the control of the OCXO designer. You can
improve the thermal path (speed wise) by moving things around. Yes, this has 
other
impacts that may or may not be an improvement in other areas.

Bob

> On Jun 12, 2022, at 5:34 AM, Louis Taber via time-nuts 
>  wrote:
> 
> I have been of the impression for years now that most "better" crystals are
> in a vacuum.  And the electrical and mechanical connections to the quartz
> itself place as little mechanical load on the crystal as possible.
> Thermal conductivity from the oven to the crystal itself would be both
> hard to model and hard to speed up.
> 
> IR transmission of energy to the crystal also seems problematic considering
> the IR transmission of quartz and the IR reflectivity of gold
> contact plating.
> 
> Is any of this an issue?
> 
>  - Louis
> 
> On Fri, Jun 10, 2022 at 9:53 PM Bob kb8tq via time-nuts <
> time-nuts@lists.febo.com> wrote:
> 
>> Hi
>> 
>>> On Jun 10, 2022, at 2:38 PM, Lux, Jim via time-nuts <
>> time-nuts@lists.febo.com> wrote:
>>> 
>>> On 6/10/22 1:57 PM, Dr. David Kirkby wrote:
>>>> On Fri, 10 Jun 2022 at 17:39, Lux, Jim via time-nuts <
>> time-nuts@lists.febo.com> wrote:
>>>> 
>>>>   On the subject of rapid warm up. I suppose if you had a need, one
>>>>   could
>>>>   dump as much power as you need into the heater. Turn on oscillator,
>>>>   lights in room dim for a few moments.
>>>> 
>>>> 
>>>> Is that not likely to damage a crystal? Different parts of the crystal
>> and likely to be at significantly different temperatures at the same time,
>> putting a lot of stress on the crystal due to a thermal gradient. It's
>> probably a bit academic, as nobody is going to make an oven that heats up
>> in fractions of a second, but if one did, I suspect it might not do the
>> crystal a lot of good. This is only an educated guess - I don't have
>> anything to back it up.
>>> Oh, it would be disastrous, although quartz is pretty strong, all the
>> rest of the mounting components might not be.
>> 
>> Indeed, breaking a quartz blank via thermal stress would be very hard to
>> do.
>> The “rest of the parts” actually are pretty durable as well. Most of it is
>> metal and
>> it is quite able to handle thermal issues.
>> 
>> The big issue in a fast warm up AT turned out to be designing the heater
>> and the
>> mount to get the energy to the blank quickly….. If you use a small enough
>> package
>> and blank, the amount of power turns out to be surprisingly small.
>> 
>> If you want to go bonkers, you mount the heaters *inside* the crystal
>> package. This
>> does indeed create some issues in various areas.
>> 
>> Bob
>> 
>>>> 
>>>> At the other extreme,  would there be any advantage in actually heating
>> the crystal very slowly, over the course of an hour/day/week, so the
>> temperature gradient across the crystal is very small? Of course, if an
>> oven took ages to reach the correct temperature, it would be inconvenient
>> for most applications, but for some applications, the advantages might
>> outweigh the disadvantages. Of course, if one does this, I suspect one
>> would have to cool the crystal slowly too to prevent a significant thermal
>> gradient across the crystal.
>>>> 
>>>> I know it's a bit different, but I have a 600 mm f4 Nikon camera lens.
>> I was told that Nikon cools the front element over a period of 6 months to
>> reduce stresses in the glass.
>>> 
>>> Big glass mirrors for telescopes do the same.
>>> 
>>> 
>>> 
>>>> 
>>>> Dave
>>> 
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

> On Jun 10, 2022, at 2:38 PM, Lux, Jim via time-nuts 
>  wrote:
> 
> On 6/10/22 1:57 PM, Dr. David Kirkby wrote:
>> On Fri, 10 Jun 2022 at 17:39, Lux, Jim via time-nuts 
>>  wrote:
>> 
>>On the subject of rapid warm up. I suppose if you had a need, one
>>could
>>dump as much power as you need into the heater. Turn on oscillator,
>>lights in room dim for a few moments.
>> 
>> 
>> Is that not likely to damage a crystal? Different parts of the crystal and 
>> likely to be at significantly different temperatures at the same time, 
>> putting a lot of stress on the crystal due to a thermal gradient. It's 
>> probably a bit academic, as nobody is going to make an oven that heats up in 
>> fractions of a second, but if one did, I suspect it might not do the crystal 
>> a lot of good. This is only an educated guess - I don't have anything to 
>> back it up.
> Oh, it would be disastrous, although quartz is pretty strong, all the rest of 
> the mounting components might not be.

Indeed, breaking a quartz blank via thermal stress would be very hard to do.
The “rest of the parts” actually are pretty durable as well. Most of it is 
metal and
it is quite able to handle thermal issues. 

The big issue in a fast warm up AT turned out to be designing the heater and 
the 
mount to get the energy to the blank quickly….. If you use a small enough 
package
and blank, the amount of power turns out to be surprisingly small. 

If you want to go bonkers, you mount the heaters *inside* the crystal package. 
This
does indeed create some issues in various areas. 

Bob

>> 
>> At the other extreme,  would there be any advantage in actually heating the 
>> crystal very slowly, over the course of an hour/day/week, so the temperature 
>> gradient across the crystal is very small? Of course, if an oven took ages 
>> to reach the correct temperature, it would be inconvenient for most 
>> applications, but for some applications, the advantages might outweigh the 
>> disadvantages. Of course, if one does this, I suspect one would have to cool 
>> the crystal slowly too to prevent a significant thermal gradient across the 
>> crystal.
>> 
>> I know it's a bit different, but I have a 600 mm f4 Nikon camera lens. I was 
>> told that Nikon cools the front element over a period of 6 months to reduce 
>> stresses in the glass.
> 
> Big glass mirrors for telescopes do the same.
> 
> 
> 
>> 
>> Dave
> 
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

Well …. folks have made AT based OCXO’s that heat up in “seconds” ( as in under
a minute ). Back in the 1980’s they stabilized to < 1x10^-7 at least as fast as 
the 
then typical SC based OCXO’s did ….  ( < 6 minutes ). Collins bought quite a 
few of 
them over the years. 

Bob

> On Jun 10, 2022, at 1:25 PM, Dr. David Kirkby via time-nuts 
>  wrote:
> 
> On Fri, 10 Jun 2022 at 17:39, Lux, Jim via time-nuts <
> time-nuts@lists.febo.com> wrote:
> 
>> 
>> On the subject of rapid warm up. I suppose if you had a need, one could
>> dump as much power as you need into the heater. Turn on oscillator,
>> lights in room dim for a few moments.
>> 
> 
> Is that not likely to damage a crystal? Different parts of the crystal and
> likely to be at significantly different temperatures at the same time,
> putting a lot of stress on the crystal due to a thermal gradient. It's
> probably a bit academic, as nobody is going to make an oven that heats up
> in fractions of a second, but if one did, I suspect it might not do the
> crystal a lot of good. This is only an educated guess - I don't have
> anything to back it up.
> 
> At the other extreme,  would there be any advantage in actually heating the
> crystal very slowly, over the course of an hour/day/week, so the
> temperature gradient across the crystal is very small? Of course, if an
> oven took ages to reach the correct temperature, it would be inconvenient
> for most applications, but for some applications, the advantages might
> outweigh the disadvantages. Of course, if one does this, I suspect one
> would have to cool the crystal slowly too to prevent a significant thermal
> gradient across the crystal.
> 
> I know it's a bit different, but I have a 600 mm f4 Nikon camera lens. I
> was told that Nikon cools the front element over a period of 6 months to
> reduce stresses in the glass.
> 
> Dave
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

There happen to be *some* AT cut based OCXO’s that beat the typical
SC cut on warmup … just saying …. :)

Bob

> On Jun 9, 2022, at 9:03 AM, Bruce Hunter via time-nuts 
>  wrote:
> 
> My only experience with SC-cut crystals is that time base oscillators in 
> later EIP/Phase-Matrix counters with SC-cut crystal oscillators seem to 
> warm-up from a cold start and stabilize more quickly than earlier AT-cut 
> versions.  I was surprised to see from Jeff Cartwright's paper that SC-cut 
> crystals operate at higher temperatures.  Could this rapid warm up 
> characteristic be attributable to oven design rather than crystal cut?
> Bruce
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

Lower turning point has been done, both with AT’s (back in ~ the 1950’s) and
with SC’s. Neither one showed any significant benefit. 

Taking a crystal down to sub 20K sort of temps does ramp up the Q. The gotcha
is that the frequency vs temp curve is so steep that very minor temperature 
variations
utterly trash the stability of the device. 

Bob

> On Jun 8, 2022, at 1:49 PM, Gerhard Hoffmann via time-nuts 
>  wrote:
> 
> Am 2022-06-08 21:53, schrieb Tom Van Baak:
>> Would it be advantageous, then, to run a high-performance laboratory
>> oscillator at its lower turnover point? Or at -78 C (CO2) or 77 K
>> (liquid Nitrogen)?
> 
> I have no idea about the crystal itself. Maybe Bernd or the SC (SantaClara)
> veterans can help?
> 
> When I measured the Q of the recovered SC crystal from that Morion MV89A,
> there was not much of a difference in the wanted resonance between room
> temperature and +89°C. I think I have published the data here a year ago.
> My deep freezer in the basement can do -36°C, but the VNA is so heavy...
> 
> Infineon boasted that their SIGET transistors work nicely at a few Kelvin,
> so it would probably not fail for semiconductor availability (BFP640 & 
> friends).
> OTOH, Ulrich Rohde wrote that the noise figure of the sustaining amplifier 
> would
> take a hit under large signal conditions, but I don't know hard numbers.
> That would not disappear.
> 
> But then, in a Driscoll for example, you can give the 2 transistors enough
> current so they run class A and do the little bit of limiting on the output 
> side
> with Schottkys. For the amplifier, that is not large signal.
> 
> That might be different for an amplifier in Lee-Hajimiri style.
> This is Dirac pulse excitation at the peak of the cycle to avoid phase 
> modulation,
> that is optimized for mixing up 1/f noise.  :-)
> 
> Anyway, with a noise figure of the sustaining amplifier of a dB or even a few,
> there is no game changer to be expected from cooling.
> 
> Whispering gallery saphire, anyone? I was at the precious stones museum
> in Idar-Oberstein here in the 'hood and saw all these huge saphires.
> I left with the head full of ideas...
> 
> Cheers, Gerhard
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

I would be careful with that paper since part of what it says is not 
(in general) correct.

Bob

> On Jun 8, 2022, at 1:22 PM, Ross P via time-nuts  
> wrote:
> 
> Hi,Thank you very much, this paper answered some questions.
> 
>On Wednesday, June 8, 2022 at 12:38:04 PM PDT, Hans-Georg Lehnard via 
> time-nuts  wrote:  
> 
> Hi, 
> 
> read this paper from Connor-Winfield about differences AT/SC cuts. 
> 
> http://www.conwin.com/pdfs/at_or_sc_for_ocxo.pdf  
> 
> Am 2022-06-08 04:04, schrieb Ross P via time-nuts:
> 
>> Hello,My first post.I have created a 64-bit frequency counter, 15.9 digits 
>> after converting to floating point. 
>> Oscillator random walk is +- 0.01 ppm with an SC cut crystal at 10 Hz 
>> filtered, and 0.1 ppm with at cut.Is it the crystal or the oscillator 
>> electronics (inside a can) that determines the noise?The oscillators I am 
>> using are 1 double oven SC 10 MHz vs 1 single oven AT cut 10 MHz in one 
>> test,and 2 generic crystal oscillators (on a Terasic DE1 cyclone II FPGA 
>> board) for the other test.I assume the single oven oscillator will have 
>> better stability than commodity oscillators.I am able to chart random walk 
>> at up to a few thousand samples per second at full double 
>> precisionresolution, and FFT shows some alien tones in the walk pattern that 
>> come and go suddenly, I thinkdue to oscillating mode changes in the 
>> oscillator itself, mostly show in the commodity crystals.My question is: is 
>> the SC quartz the most stable for random walk.I would like to know if such a 
>> frequency counter / alien to detector is useful enough to be producedfor 
>> sale? It would require at least 3 separate frequencies of
> refer
>> ence time standards and > 50Klogic elements in the FPGA for 3 cross coupled 
>> monitors to cover a range of 0 to 50 MHz. 
>> Quite a risk if no one needs it. 3 separate high stability reference 
>> oscillators are expensive.rp
>> 
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

Well ….. 

You can bash both AT’s and SC’s a *lot* harder than you might think. Both
will suffer quite a bit in terms of ADEV when you do. 

Since the AT likely has a lower resistance (by quite a bit) than the SC, the 
loop
current ( and thus the drive into the buffer) may not be as far different on the
two as you would guess.

Bob

> On Jun 8, 2022, at 10:42 AM, Gerhard Hoffmann via time-nuts 
>  wrote:
> 
> Am 2022-06-08 13:27, schrieb Magnus Danielson via time-nuts:
> 
>> As far as I remember and know, you can achieve about the same
>> phase-noise properties as you hit about the same bandwidth from the Q,
>> and noise contribution is about the same. So, it boils down to do the
>> supporting amplifier well.
> 
> But SC can tolerate more power, so you may get more distance to the
> thermal noise floor.
> 
> cheers, Gerhard
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[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

The stability depends on a long list of things. Since you can get higher
Q with an AT than you can with an SC, if Q was all that mattered, the AT
would be the king of the hill. 

The oscillator circuit matters, but different parts of it matter in different 
ways.
The things you might do for low phase noise at a 100KHz offset might be 
a bad idea if very good ADEV at 100 seconds was the target. 

Tuning any high Q circuit very far off frequency probably is not a great idea.
Keeping all of the “optimizations” on target over a wide pull range is not at
all simple. 

If you are designing an OCXO from scratch, there is a lot to learn and hundreds
of papers out there to get you started. If you are buying one, things are a bit
more simple. You look at the spec sheet and decide if it’s going to do the job
or not. Worst case, you buy a couple and test them. 

Bob

> On Jun 8, 2022, at 10:01 AM, Ross P via time-nuts  
> wrote:
> 
> Hi,So, highest short term stability depends on the Q of the crystal and 
> quality of the feedback circuit. In that case, an AT-cut with a low noise 
> feedback amplifier will be as good as an SC-cut with the same amp. Does 
> pulling the oscillator affect the short term walk?rp
> 
>On Wednesday, June 8, 2022 at 10:44:24 AM PDT, Magnus Danielson via 
> time-nuts  wrote:  
> 
> Hi,
> 
> I agree in general. However, I do see that other work to get good 
> resulst have been done when SC-cut is considered, so rather than SC-cut 
> as a cut is better, it becomes somewhat of a tell-tale of that other 
> work being done properly. I.e. it is meaningless to take the step to 
> SC-cut when other defects dominate so the SC-cut properties only makes 
> things more expensive than the AT-cut.
> 
> As far as I remember and know, you can achieve about the same 
> phase-noise properties as you hit about the same bandwidth from the Q, 
> and noise contribution is about the same. So, it boils down to do the 
> supporting amplifier well.
> 
> Cheers,
> Magnus
> 
> On 2022-06-08 06:27, Bob kb8tq via time-nuts wrote:
>> Hi
>> 
>> Simple answer is: no.
>> 
>> More complete answer is: no
>> 
>> There is a lot more to stability than just the crystal cut. Having this or 
>> that cut is
>> in no way a guarantee that the result is “better” than some other cut. 
>> Indeed there
>> are more exotic cuts than the SC that improve on this or that. There are 
>> also mounting
>> / fabrication techniques that improve on this or that, regardless of cut.
>> 
>> All that said, the “typical” SC cut based OCXO is likely newer than an AT or 
>> BT cut
>> alternative. Various improvements here or there are likely to make it a bit 
>> better than
>> the other examples …. ( but not always )
>> 
>> Bob
>> 
>>> On Jun 7, 2022, at 6:04 PM, Ross P via time-nuts  
>>> wrote:
>>> 
>>> Hello,My first post.I have created a 64-bit frequency counter, 15.9 digits 
>>> after converting to floating point.
>>> Oscillator random walk is +- 0.01 ppm with an SC cut crystal at 10 Hz 
>>> filtered, and 0.1 ppm with at cut.Is it the crystal or the oscillator 
>>> electronics (inside a can) that determines the noise?The oscillators I am 
>>> using are 1 double oven SC 10 MHz vs 1 single oven AT cut 10 MHz in one 
>>> test,and 2 generic crystal oscillators (on a Terasic DE1 cyclone II FPGA 
>>> board) for the other test.I assume the single oven oscillator will have 
>>> better stability than commodity oscillators.I am able to chart random walk 
>>> at up to a few thousand samples per second at full double 
>>> precisionresolution, and FFT shows some alien tones in the walk pattern 
>>> that come and go suddenly, I thinkdue to oscillating mode changes in the 
>>> oscillator itself, mostly show in the commodity crystals.My question is: is 
>>> the SC quartz the most stable for random walk.I would like to know if such 
>>> a frequency counter / alien to detector is useful enough to be producedfor 
>>> sale? It would require at least 3 separate frequencies of refer
>>> ence time standards and > 50Klogic elements in the FPGA for 3 cross coupled 
>>> monitors to cover a range of 0 to 50 MHz.
>>> Quite a risk if no one needs it. 3 separate high stability reference 
>>> oscillators are expensive.rp
>>> 
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>&

[time-nuts] Re: Is SC the most stable cut for lowest phase noise?

[Bob kb8tq via time-nuts]

Hi

Simple answer is: no.

More complete answer is: no

There is a lot more to stability than just the crystal cut. Having this or that 
cut is 
in no way a guarantee that the result is “better” than some other cut. Indeed 
there
are more exotic cuts than the SC that improve on this or that. There are also 
mounting
/ fabrication techniques that improve on this or that, regardless of cut. 

All that said, the “typical” SC cut based OCXO is likely newer than an AT or BT 
cut
alternative. Various improvements here or there are likely to make it a bit 
better than
the other examples …. ( but not always )

Bob

> On Jun 7, 2022, at 6:04 PM, Ross P via time-nuts  
> wrote:
> 
> Hello,My first post.I have created a 64-bit frequency counter, 15.9 digits 
> after converting to floating point. 
> Oscillator random walk is +- 0.01 ppm with an SC cut crystal at 10 Hz 
> filtered, and 0.1 ppm with at cut.Is it the crystal or the oscillator 
> electronics (inside a can) that determines the noise?The oscillators I am 
> using are 1 double oven SC 10 MHz vs 1 single oven AT cut 10 MHz in one 
> test,and 2 generic crystal oscillators (on a Terasic DE1 cyclone II FPGA 
> board) for the other test.I assume the single oven oscillator will have 
> better stability than commodity oscillators.I am able to chart random walk at 
> up to a few thousand samples per second at full double precisionresolution, 
> and FFT shows some alien tones in the walk pattern that come and go suddenly, 
> I thinkdue to oscillating mode changes in the oscillator itself, mostly show 
> in the commodity crystals.My question is: is the SC quartz the most stable 
> for random walk.I would like to know if such a frequency counter / alien to 
> detector is useful enough to be producedfor sale? It would require at least 3 
> separate frequencies of refer
> ence time standards and > 50Klogic elements in the FPGA for 3 cross coupled 
> monitors to cover a range of 0 to 50 MHz. 
> Quite a risk if no one needs it. 3 separate high stability reference 
> oscillators are expensive.rp
> 
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[time-nuts] Re: Identifying GPSDO phase disturbers

[Bob kb8tq via time-nuts]

Hi

It’s a good bet that the TCXO has an AT cut crystal in it. That would 
give it a 1 to 2 ppb / G sensitivity for typical blanks mounted in a 
normal fashion. 

Quantifying the shock delivered by a “tap” on a part can be exciting. 
It can be done, but the gear to do it properly does cost more than most
of us would like to spend. You can get some pretty large G levels for
very short durations ( 100 G’s for 1 ms maybe ). 

Combine the two and you get a very measurable change in the part
from some pretty simple messing around. 

Bob

> On Jun 6, 2022, at 9:05 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> During measurement  of a  GPSDO there was some concern about very short term 
> phase stability. E.g. for tau between 0.001 and 1 second. It proved to be 
> possible to measure the stability for tau larger than 0.1 s using a frequency 
> counter but neither the counter (limited accuracy for very short tau) nor 
> Timelab (shortest tau was 0.02 s) where able to reach a tau of 0.01 s.
> Looking at the old HP/Agilent application notes a phase detector approach was 
> selected.
> The output of the GPSDO was send to the RF port of a mixer. The LO port was 
> connected to the output of a VC-TCXO and the IF port output was low pass 
> filtered (to remove the 10MHz and higher) and added to the Vtune to the 
> VC-TCXO. Course tuning of the VC-TCXO was done using a 10 turn potmeter 
> supplied from a very stable linear supply.
> It proved to be possible to set the V-tune with the potmeter such that the 
> GPSDO and VC-TCXO frequencies where in phase and the loop locked.
> Using a dual input frequency counter the ratio of the GPSDO and VC_TXCO was 
> measured to confirm they where in lock.
> An oscilloscope with FFT was also connected to the LPF output to monitor 
> short term phase disturbances. No high frequency (above 10Hz)  components 
> where observed in the FFT suggesting the initial concern was not justified
> Three main phase disturbances where observed.
> 1: The GPSDO was in phase lock with the GPS PPS and every time the tuning DAC 
> was updated a change in frequency resulting in a change in angle on the scope 
> of the mixer output was observed. These changes where also visible on the 
> frequency counter
> 2: Any temperature changed caused strong phase fluctuations. Even tough the 
> GPSDO uses a TCXO there is still a large temperature sensitivity. Thermal 
> isolation (adding some towels) helped to remove fast temperature fluctuations.
> 3: Mechanical shock caused clearly visible phase variations. The VC-TCXO 
> acted as a sensitive microphone and, to a lesser extend, also the TCXO of the 
> GPSDO. Tapping on the workbench with one finger was visible. The net effect 
> of the mechanical shock was about zero phase change which made it difficult 
> to see on the frequency counter with 0.1 s gate time but the higher BW of the 
> phase detector allowed to observe this. It is yet unclear how to isolate the 
> TCXO in the GPSDO from mechanical shock
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[time-nuts] Re: Turning off display on HP 58503 A or B

[Bob kb8tq via time-nuts]

Hi

Both control loops in the 3801 OCXO’s are thermistor based. If the outside loop 
goes nuts, the device might survive. If the inside loop shorts out, the normal 
result is a melt down of the OCXO innards. 

Bob

> On Jun 2, 2022, at 9:55 AM, K5jv via time-nuts  
> wrote:
> 
> I was recently evolved in a discussion about the HP 58503 option 001 display. 
>  The general idea is that there is some command to tune the display on or 
> off.  A few ideas were offered but no one had anything except "they heard 
> about it."   I have found nothing on the subject from old HP literature.  
> Comments would be appreciated.  I am not talking about turning the power off 
> to the DC/DC converter.
> 
> On a different subject:  A couple of month back I got some strong criticism 
> about suggesting that the oven in the Z3801A was a potential source of 
> problems.   I just got another Z3801A in for repair.  Same symptoms as 
> before;  appears to turn on normally but after a while goes into holdover and 
> stay there,  same LH flat lines.  Once again the problem turned out to be the 
> oven.  It was unusually hot to the touch.  Changing the oven solved the 
> problem.  Now my question.  I have never opened one of these oven, so have no 
> real idea exactly what is inside.  I would  suspect that one of  the oven's 
> thermostats is stuck in the "on" position.  Since it is a "double oven",  
> there could be two thermostats?   If these are mechanical contacts, as one 
> might expect fro the 1995 era, can they be cleaned or replaced?   Comments 
> from anyone who has actually seen inside one of these ovens would be 
> appreciated.
> 
> 
> 73 de Lon, K5JV
> 281-795-1335
> 
> 
> [a96323c7-47c0-46b2-8018-f039618476d5]
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[time-nuts] Re: Realtime comparing PPS of 3 GPS

[Bob kb8tq via time-nuts]

Hi

The variation you see is dependent on a number of things. One of them is space 
weather. If you do your run during a very active period ( typically peak sun 
spots)
you may see some very dramatic swings on a single band device. 

Bob

> On May 30, 2022, at 3:00 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> Further evaluation did shown the time differences between the 3 GPS modules 
> was due to difference in the trigger level setting of the timer/counter and 
> difference in length of GPS antenna cables.
> After removal of the phase drift due to Rb frequency offset the attached 
> image shows the phase differences of the 3 modules versus a Rb reference.
> The two ATGM modules are very consistent over a 2.8 hours period. The NEO-7M 
> varies wildly  with phase errors above 100 ns. Possibly due to a somewhat 
> less optimal antenna position.
> It seems phase variations over time in the order of 10-20 ns are indeed 
> unavoidable, even with a good antenna.
> Erik.
> <3GPS_phase_difference.png>___
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[time-nuts] Re: Ublox M6T -M8T

[Bob kb8tq via time-nuts]

Hi

There are a number of threads on the uBlox receivers. 

Quick summary: All of the outputs from the typical examples are run
with a “drop a pulse / add a pulse” approach to steer them to the 
correct frequency. If you are after a “clean” output for some sort of
gear (like a synthesizer) … not so much. If further division for a timing 
process, they may be useful.

The drop / add happens on a second to second basis. You get a 
“step” in the process each time a PPS “happens”. This results in 
a bit of excitement when trying to measure just how good (or bad) 
the output is. 

Bob

> On May 28, 2022, at 9:05 AM, R Putz via time-nuts 
>  wrote:
> 
> Has anyone done anything with the Ublox GPS timing receivers? As it appears 
> the 
> Navman with the 10khz outputs seem to be drying up, I'm wondering about the 
> Time Pulse 2 output being set to 10 Khz or 100 Khz. Thoughts anyone?
> Rich
> W9ENG
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[time-nuts] Re: Optimizing GPSDO for phase stability

[Bob kb8tq via time-nuts]

Hi

What is the “customer” after? 

For a PPS, it could be the offset from UTC. This gets into GPS to UTC and then 
into
GPS master to local GPS pulse. Bottom line usually is that the raw GPS pulse is 
the answer to “close to UTC”. Equally it could be a PPS used for metrology ( = 
ADEV 
measurement). Then you want the lowest ADEV PPS. The answer here is a PPS 
divided
off the local oscillator with nothing else done to it. 

Different customers, different needs, different answers. The same applies to how
the local oscillator is disciplined. 

What to do (without building a dozen different designs?) ….. give the customer 
a 
software setting that lets them pick what they get. Write up a couple dozen 
pages
on why you would use one or the other. Yes, the software switch takes about two 
minutes to code. The pages of “yack” may take a couple weeks to fully sort out. 

Bob

> On May 27, 2022, at 8:02 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> The GPSDO/Timer/Counter I'm building also is intended to have a stabilized 
> PPS output (so with GPS jitter removed).
> The output PPS is created by multiplying/dividing the 10MHz of a disciplined 
> TCXO up and down to 1 Hz using a PLL and a divide by 2e8. No SW or re-timing 
> involved.
> The 1 PPS output is phase synchronized with the PPS using a SW control loop 
> and thus should be a good basis for experiments that require a time pulse 
> that is stable and GPS time correct.
> As I have no clue how to specify or evaluate the performance of such a PPS 
> output I've done some experiments.
> In the first attached graph you can see the ADEV of the GPS PPS (PPS - Rb) 
> and the 1 PPS output with three different control parameters (Tick - RB)
> As I found it difficult to understand what the ADEV plot in practice means 
> for the output phase stability I also added the Time Deviation plot as I'm 
> assuming this gives information on the phase error versus the time scale of 
> observation.
> Lastly a plot is added showing the Phase Difference. All plots where created 
> using the linear residue as the Rb used as reference is a bit out of tune.
> Also the TIM files are attached
> The "PPS - RB" and "Tick - RB Kp=0.04" where measured simultaneously and 
> should show the extend to which the GPS PPS is actually drifting in phase 
> versus the Rb and how this impacts the output phase of the stabilized output 
> PPS.
> My conclusion is that a higher then expected Kp of 0.1 gives the most stable 
> output phase performance where the best frequency performance is realized 
> with a Kp = 0.04
> I welcome feedback on the interpretation of these measurements and the 
> application of output phase stabilization.
>  Rb.tim> Kp=0.04.tim>___
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[time-nuts] Re: measuring tiny devices

[Bob kb8tq via time-nuts]

Hi

That’s one of the methods. There are others for various filter topologies. Some 
are
more practical than others …

Bob

> On May 26, 2022, at 9:27 AM, John Lofgren  
> wrote:
> 
> Bob,
> 
> You may be thinking of Dishal's method.
> < 
> https://www.johansontechnology.com/dishal-bandpass-filter-tuning-using-lasertrim-chip-caps>
> 
> -John
> 
> -----Original Message-
> From: Bob kb8tq via time-nuts 
> Sent: Thursday, May 26, 2022 10:18 AM
> To: Discussion of precise time and frequency measurement 
> 
> Cc: Bob kb8tq 
> Subject: [time-nuts] Re: measuring tiny devices
> 
> EXTERNAL EMAIL: Be careful with attachments and links.
> 
> Hi
> 
> The real answer to the problem is to dig into the bowels of 1940’s electronic 
> craft.
> There are various methods for setting up an L/C filter. You short this / open 
> that sweep to find a dip or a peak. You move it to the “right” place. Just 
> what you do depends very much on the filter design. Many L/C’s got done this 
> way or that way simply because they would fit a known alignment method.
> 
> While it all sounds very cumbersome and obscure it actually isn’t. Long ago I 
> stumbled upon a gal setting up very complex L/C IF filters this way. The 
> display gyrated this way and that way as she did this or that. I don’t think 
> it took her more than a minute to get the whole thing set up….. to this day, 
> I’m amazed by how fast she was.
> 
> Do I have any useful links to actually read up on  this magic? … sorry about 
> that.
> 
> Bob
> 
>> On May 26, 2022, at 4:58 AM, Lux, Jim via time-nuts 
>>  wrote:
>> 
>> On 5/25/22 3:16 PM, ed breya via time-nuts wrote:
>>> Thanks Mike, for info on LCR alternatives. It's good to know of others out 
>>> there, if needed. I have an HP4276A and HP4271A. The 4276A is the main 
>>> workhorse for all part checking, since it has a wide range of LCZ, although 
>>> limited frequency coverage (100 Hz - 20 kHz). The 4271A is 1 MHz only, and 
>>> good for smaller and RF parts, but very limited upper LCR ranges. I think 
>>> it works, so I can use it if needed, but would have to check it out and 
>>> build an official lead set for it. I recall working on it a few years ago 
>>> to fix some flakiness in the controls, so not 100% sure of its present 
>>> condition.
>>> 
>>> The main difficulty I've found in measuring small chokes is more of 
>>> probing/connection problem rather than instrument limitation. For most 
>>> things, I use a ground reference converter that I built for the 4276A many 
>>> years ago. It allows ground-referenced measurements, so the DUT doesn't 
>>> have to float inside the measuring bridge. The four-wire arrangement is 
>>> extended (in modified form) all the way to a small alligator clip ground, 
>>> and a probe tip, for DUT connection, so there is some residual L in the 
>>> clip and the probe tip, which causes some variable error, especially in 
>>> attaching to very small parts and leads. When you add in the variable 
>>> contact resistance too, it gets worse. Imagine holding a small RF can 
>>> (about a 1/2 inch cube) between your fingers, with a little clip sort of 
>>> hanging from one lead, and pressing the end of the probe tip against the 
>>> other lead. All the while, there's the variable contact forces, and effects 
>>> from the relative positions of all the pieces and fingers, and the stray C 
>>> from the coil to the can to the fingers. I have pretty good dexterity, and 
>>> have managed to make these measurements holding all this stuff in one hand, 
>>> while tweaking the tuning slug with the other.
>>> 
>>> I had planned on making other accessories like another clip lead to go in 
>>> place of the probe tip, but not yet built. I also have the official 
>>> Kelvin-style lead set that came with the unit, so that's an option that 
>>> would provide much better accuracy and consistency, but the clips are 
>>> fairly large and hard to fit in tight situations, and the DUT must float. 
>>> Anyway, I can make all sorts of improvements in holding parts and hookup, 
>>> but usually I just clip and poke and try to get close enough - especially 
>>> when I have to check a lot of parts, quickly.
>>> 
>>> The other problem is that the 4276A is near its limit for getting 
>>> measurements below 1 uH, with only two digits left for nH. The 4271A would 
>>> be much better for this, with 1 nH vs 10 nH resolution.
>>> 
>>> If I get in a situation where I need to do a lot of this (if I sh

[time-nuts] Re: measuring tiny devices

[Bob kb8tq via time-nuts]

Hi

The real answer to the problem is to dig into the bowels of 1940’s electronic 
craft.
There are various methods for setting up an L/C filter. You short this / open 
that sweep
to find a dip or a peak. You move it to the “right” place. Just what you do 
depends
very much on the filter design. Many L/C’s got done this way or that way simply
because they would fit a known alignment method.

While it all sounds very cumbersome and obscure it actually isn’t. Long ago I 
stumbled
upon a gal setting up very complex L/C IF filters this way. The display gyrated 
this way
and that way as she did this or that. I don’t think it took her more than a 
minute to get 
the whole thing set up….. to this day, I’m amazed by how fast she was.

Do I have any useful links to actually read up on  this magic? … sorry about 
that.

Bob

> On May 26, 2022, at 4:58 AM, Lux, Jim via time-nuts 
>  wrote:
> 
> On 5/25/22 3:16 PM, ed breya via time-nuts wrote:
>> Thanks Mike, for info on LCR alternatives. It's good to know of others out 
>> there, if needed. I have an HP4276A and HP4271A. The 4276A is the main 
>> workhorse for all part checking, since it has a wide range of LCZ, although 
>> limited frequency coverage (100 Hz - 20 kHz). The 4271A is 1 MHz only, and 
>> good for smaller and RF parts, but very limited upper LCR ranges. I think it 
>> works, so I can use it if needed, but would have to check it out and build 
>> an official lead set for it. I recall working on it a few years ago to fix 
>> some flakiness in the controls, so not 100% sure of its present condition.
>> 
>> The main difficulty I've found in measuring small chokes is more of 
>> probing/connection problem rather than instrument limitation. For most 
>> things, I use a ground reference converter that I built for the 4276A many 
>> years ago. It allows ground-referenced measurements, so the DUT doesn't have 
>> to float inside the measuring bridge. The four-wire arrangement is extended 
>> (in modified form) all the way to a small alligator clip ground, and a probe 
>> tip, for DUT connection, so there is some residual L in the clip and the 
>> probe tip, which causes some variable error, especially in attaching to very 
>> small parts and leads. When you add in the variable contact resistance too, 
>> it gets worse. Imagine holding a small RF can (about a 1/2 inch cube) 
>> between your fingers, with a little clip sort of hanging from one lead, and 
>> pressing the end of the probe tip against the other lead. All the while, 
>> there's the variable contact forces, and effects from the relative positions 
>> of all the pieces and fingers, and the stray C from the coil to the can to 
>> the fingers. I have pretty good dexterity, and have managed to make these 
>> measurements holding all this stuff in one hand, while tweaking the tuning 
>> slug with the other.
>> 
>> I had planned on making other accessories like another clip lead to go in 
>> place of the probe tip, but not yet built. I also have the official 
>> Kelvin-style lead set that came with the unit, so that's an option that 
>> would provide much better accuracy and consistency, but the clips are fairly 
>> large and hard to fit in tight situations, and the DUT must float. Anyway, I 
>> can make all sorts of improvements in holding parts and hookup, but usually 
>> I just clip and poke and try to get close enough - especially when I have to 
>> check a lot of parts, quickly.
>> 
>> The other problem is that the 4276A is near its limit for getting 
>> measurements below 1 uH, with only two digits left for nH. The 4271A would 
>> be much better for this, with 1 nH vs 10 nH resolution.
>> 
>> If I get in a situation where I need to do a lot of this (if I should get 
>> filter madness, for instance), then I'll have to improve the tools and 
>> methods, but I'm OK for now, having slogged through it this time. 
> 
> 
> You might check out the NanoVNA - people have made a variety of novel 
> fixtures for measuring small parts (i.e. 0604 SMTs)
> 
> It certainly has the measurement frequency range you need. The trick is 
> figuring out whether you want to do a series or shunt measurement, and that 
> sort of depends on the reactance of your device at the frequency of interest.
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[time-nuts] Re: Looking for an SMA GPS signal splitter

[Bob kb8tq via time-nuts]

HI

None of them mine:

https://www.ebay.com/itm/144026433966 

https://www.ebay.com/itm/201036132548 

https://www.ebay.com/itm/112402977059 
https://www.ebay.com/itm/233189112130 

https://www.ebay.com/itm/293666952089 

https://www.ebay.com/itm/403461970783 

https://www.ebay.com/itm/132126337610 


(and on and on ….)

The two way’s are tossed in since I’ve found it better to put a 2 way up front 
and then a couple of 2’s or 4’s ( or 16’s …)
after that. There’s nothing magic about it. The cabling of this or that just 
seems to be easier.

Bob




> On May 24, 2022, at 6:37 AM, John Miller via time-nuts 
>  wrote:
> 
> Hello everyone,
> I'm looking for another GPS signal splitter. My plan was to get a "generic" 
> SMA splitter/combiner and a handful of SMA DC blocking adapters, but they are 
> considerably more expensive than I had expected - and they all seem to be in 
> China. If anyone (in the US) has an extra 4 or 6-way splitter w/ DC blocks 
> that they are interested in selling, please let me know!
> 
> Thanks!
> 
> 
> John
> KC1QLN
> millerjs.org
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[time-nuts] Re: Does filtering a TCXO Vtune to reduce low tau ADEV with max 50% make sense?

[Bob kb8tq via time-nuts]

Hi

The typical answer to this issue is to first try to clean up the supply to the
DAC. If that does not fix the issue, you likely need a better DAC. Unless the
TCXO has a crazy large tune range, rational parts should be able to do the job.
If the TCXO has a crazy large range then maybe a different TCXO is the answer.

You can measure this or that to confirm / deny that the problem exists here or 
there in the circuit. An audio analyzer that gets down to a bit under 1 Hz 
should
be able to tell you what’s what. Various sound card based approaches likely 
would be “good enough” ( yes, ground loops will be fun to take care of ….)

Bob

> On May 20, 2022, at 9:26 AM, Erik Kaashoek via time-nuts 
>  wrote:
> 
> During testing a TCXO using a correct Vtune for 10MHz output, the ADEV at 
> tau=1 second  was measured at 1.08E-10 (see attached plot[1], trace 
> DAC=normal)
> When the DAC controlling the Vtune was forced to zero volt output the ADEV at 
> tau=1 s reduced to 5.43E-11 (see plot, trace DAC=0)
> This clearly hinted at some noise present on the DAC output.
> Adding substantial filtering between the DAC and the Vtune input of the TCXO 
> and setting to 10MHz resulted in an ADEV at tau=1 s of 7.28E-11 (see plot, 
> trace DAC filtered)
> All measurements where done with the controller for the Vtune disabled.
> The plot contains error bars.
> 
> The extra filtering required for the reduction creates a lag of about 10 
> seconds in controlling the TCXO and the controller without the filtering was 
> nicely able to correct fairly quickly random walks so I'm a bit worried about 
> the problems this 10 seconds extra lag may create in tuning the controller 
> loop.
> 
> Is a reduction of the ADEV of a 10MHz reference output with 25% (or at most 
> 50%) for low tau in practice of any relevance if this may cause an increase 
> in ADEV for larger tau as the controller may have more difficulty to 
> correctly adjust Vtune to correct random walks?
> 
> [1] Plot: 
> http://athome.kaashoek.com/time-nuts/DAC_Filtering.png___
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[time-nuts] Re: Effect of temperature on cheap puck style GNSS antennas?

[Bob kb8tq]

Hi


> On May 12, 2022, at 3:21 AM, Lux, Jim  wrote:
> 
> On 5/11/22 11:50 PM, Matthias Welwarsky wrote:
>> Dear list members,
>> 
>> My DIY GPSDO has a rather well defined dependence to the environmental
>> temperature, which correlates almost linearly with a frequency shift of the
>> OCXO. However, at times I see the error against the GNSS reference increasing
>> with its case temperature not warranting such effect.
>> 
>> My antenna is one of those cheap, magnetic, active antennas you'd put on a 
>> car
>> roof. It's facing south and has full exposure to the sun, obviously.
>> 
>> During sunrise I see the TIC error increasing 20ns-30ns over lets say 2000
>> seconds. The GPSDO case temperature rises, too, during that time as the room
>> temperature increases, but it is only by 0.3°C.
>> 
>> I'm wondering if the temperature of the antenna, which of course rises much
>> faster than the room temperature, can have an effect of this magnitude?
> 
> 
> Very possible. I've seen fairly large changes (nanoseconds over a 0-40C temp 
> range) in delay in the LNA and bandpass filter for GNSS receivers with 
> temperature. If they're using any sort of ceramic filter or ceramic antenna, 
> then that can have a fairly large tempco in the time delay.

The ceramic typically used for antennas is unlikely to have that much change 
over any reasonable temperature range. The ceramic filters are very different
beasts …. The impact of the antenna should be down in the “couple of ns” range 
at most. 

Since this is a “who knows what” antenna, there is no way to be *sure* of what 
it’s 
doing. A properly designed small / low cost antenna should do pretty well. 

Bob

> 
> 
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[time-nuts] Re: Can the ADEV of a GPSDO output ever be lower than the minimum of the ADEV of the internal oscilator and the ADEV of the GPS PPS?

[Bob kb8tq]

HI 

> On May 10, 2022, at 5:42 AM, Attila Kinali  wrote:
> 
> On Tue, 10 May 2022 12:49:27 +0200
> Erik Kaashoek  wrote:
> 
>> Error bars like this?
> 
> Exactly...
> 
> And what you see is, that beyond ~300s the difference is not statistically
> significant anymore and beyond 700s the curves are indistinguishable.
> 
> I'm a bit surprised that a TCXO (with control loop) is good enough for 100s.
> Even if the difference to the PPS is only a factor of 1.6. 
> That's a quite decent oscillator.

or it’s in a *very* unusual temperature environment ….

Bob

> 
>   Attila Kinali
> -- 
> In science if you know what you are doing you should not be doing it.
> In engineering if you do not know what you are doing you should not be doing 
> it.
>-- Richard W. Hamming, The Art of Doing Science and Engineering
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[time-nuts] Re: OSA-5400 power transistor

[Bob kb8tq]

Hi

With 24V in, 450 ma is 10.8W. That’s a pretty hearty number for warming up
an OCXO. That power would not be going through the 18V regulator for a couple
of reasons. 

Once warmed up, 170 ma at 24V is 4.08W. That’s more than a functioning OCXO
should pull at typical lab bench temperatures. 60 ma at 24V is 1.44W. That is a 
reasonable power for a functioning OCXO on a lab bench. 

Bob

> On May 9, 2022, at 3:32 PM, Marek Doršic  wrote:
> 
> Hi Paul,
> 
>thanks for your insight. Sadly there is no oscillations on the 18V. 
> The problem must be somewhere else. What I just do not understand is, why it 
> starts working for couple of hours when I changed the Q4 PNP transistor. And 
> now all voltages seems to be fine and it did not work. I also today noticed 
> interestingly that touching the Q1 transistor metal case with a voltage probe 
> leads to huge input current changes. While heating-up, when I touch it the 
> current increases from 440mA to 460mA. When the unit was already heated, by 
> touching the transistor case the input current drops from 170mA to only 60mA. 
> I am not capable to understand the circuit right now, but for me this might 
> be the next suspicious part.
> 
> The oven heating seems to be OK. On the PCB are pins labeled with C, B, E ( 
> photo 
>  here) with 
> wires soldered leading directly to the heated core. I measured voltages on 
> these pins together with the input current. See attached graph 
> 
>  below of first 40 minutes while the unit was heating up. I think this pins 
> are the terminals of the heating transistor inside the oven and everything 
> looks good there.  0.44A@24V = 10.5W and then the current settles down at 
> about 0.17A (4W). This is quite in within spec (should be 3.5W after 2.5 
> hours warm-up).
> 
> 
> 
>   .marek
> 
>> On 9 May 2022, at 15:54, paul swed  wrote:
>> 
>> Marek
>> Thanks. I have the schematic and can now see that its a 18V regulator. So
>> thats only 3 watts. Its a classic differential regulator so it can accept a
>> wide range of transistors because the circuit has quite a bit of gain. If
>> your transistor is being destroyed then potentially there is an oscillation
>> in the circuit.
>> A scope on the +18 should tell you.
>> Other then that the current should start high at .46 amps as you mention in
>> by 20 minutes should drop down to 46 ma as a guess. If it stays high the
>> ovens overheating and as you are concerned perhaps a bad themistor.
>> Let us know how you are doing.
>> Regards
>> Paul
>> WB8TSL
>> 
>> On Mon, May 9, 2022 at 9:11 AM Marek Doršic > > wrote:
>> 
>>> Yes, it is a power transistor with heatsing.
>>> Please find attachned the attachments via dropbox
>>> 
>>> 
>>> https://www.dropbox.com/s/efzgvs2rh8c76in/Screenshot%202022-05-08%20at%2018.58.10.png?dl=0
>>>  
>>> 
>>> <
>>> https://www.dropbox.com/s/efzgvs2rh8c76in/Screenshot%202022-05-08%20at%2018.58.10.png?dl=0
>>>  
>>> 
 
>>> 
>>> https://www.dropbox.com/s/wd4yrndn4scfzov/Screenshot%202022-05-08%20at%2019.01.46.png?dl=0
>>>  
>>> 
>>> <
>>> https://www.dropbox.com/s/wd4yrndn4scfzov/Screenshot%202022-05-08%20at%2019.01.46.png?dl=0
>>>  
>>> 
 
>>> 
>>> .marek
>>> 
 On 8 May 2022, at 21:05, paul swed  wrote:
 
 Marek
 No diagram included that I can see.
 The next comment may be totally wrong since I have nothing to go on.
 If the input voltage is 24 V and the supply is 10 V reg at .48A, then
 during the initial warm up the transistor easily dissipates 6 watts. That
 would be a power transistor and some form of heat sink to keep the
>>> junction
 temperature reasonable.
 Regards
 Paul
 WB8TSL
 
 On Sun, May 8, 2022 at 2:14 PM Marek Doršic >> >> wrote:
 
> I would like to get your thoughts on my problem with OSA-5400
>>> oscillator.
> 
> I have on old unit, which is somehow broken. I was told it was
>>> overpowered
> with voltages up to 32V (standard supply voltage is 24V) and even
>>> sourced
> with reverse polarity supply power.
> 
> When I first powered it up, it draws only 2mA. I replaced what I
>>> supposed
> was a broken 10V voltage reference (how wrong I was), with a 10V zener
> diode and voilà, I had a nice steady 5MHz, 14dB signal. But only for
>>> couple
> of hours and then it died 

[time-nuts] Re: Simple simulation model for an OCXO?

[Bob kb8tq]

Hi

The most basic is the “phase pop” that is not modeled by any of the 
normal noise formulas. The further you dig in, the more you find things
that the models really don’t cover. 

Bob

> On May 4, 2022, at 11:50 AM, Attila Kinali  wrote:
> 
> Hoi Bob,
> 
> On Tue, 3 May 2022 16:23:27 -0500
> Bob kb8tq  wrote:
> 
>> The gotcha is that there are a number of very normal OCXO “behaviors” that 
>> are not
>> covered by any of the standard statistical models. 
> 
> Could you elaborate a bit on what these "normal behaviours" are?
> 
>   Attila Kinali
> 
> -- 
> The driving force behind research is the question: "Why?"
> There are things we don't understand and things we always 
> wonder about. And that's why we do research.
>   -- Kobayashi Makoto
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[time-nuts] Re: Simple simulation model for an OCXO?

[Bob kb8tq]

Hi

The gotcha is that there are a number of very normal OCXO “behaviors” that are 
not
covered by any of the standard statistical models. Coping with these issue is 
at least
as important at working with the stuff that is coved by any of the standard 
statistical 
models ….

Bob

> On May 3, 2022, at 3:57 AM, Matthias Welwarsky  wrote:
> 
> Dear all,
> 
> thanks for your kind comments, corrections and suggestions. Please forgive if 
> I don't reply to all of your comments individually. Summary response follows:
> 
> Attila - yes, I realize temperature dependence is one key parameter. I model 
> this meanwhile as a frequency shift over time.
> 
> Bob - I agree in principle, real world data is a good reality check for any 
> model, but there are only so few datasets available and most of the time they 
> don't contain associated environmental data. You get a mix of effects without 
> any chance to isolate them.
> 
> Magnus, Jim - thanks a lot. Your post encouraged me to look especially into 
> flicker noise an how to generate it in the time domain. I now use randn() and 
> a low-pass filter. Also, I think I understood now how to create phase vs 
> frequency noise.
> 
> I've some Timelab screenshots attached, ADEV and frequency plot of a data set 
> I generated with the following matlab function, plus some temperature 
> response 
> modeled outside of this function.
> 
> function [phase] = synth_osc(samples,da,wpn,wfn,fpn,ffn)
>   # low-pass butterworth filter for 1/f noise generator
>   [b,a] = butter(1, 0.1);
> 
>   # aging
>   phase = (((1:samples)/86400).^2)*da;
>   # white phase noise
>   phase += (randn(1, samples))*wpn;
>   # white frequency noise
>   phase += cumsum(randn(1, samples))*wfn;
>   # 1/f phase noise
>   phase += filter(b,a,randn(1,samples))*fpn;
>   # 1/f frequency noise
>   phase += cumsum(filter(b,a,randn(1,samples))*ffn);
> end
> 
> osc = synth_osc(40, -50e-6, 5e-11, 1e-11, 5e-11, 5e-11);
> 
> Thanks.
> 
> On Montag, 2. Mai 2022 17:12:47 CEST Matthias Welwarsky wrote:
>> Dear all,
>> 
>> I'm trying to come up with a reasonably simple model for an OCXO that I can
>> parametrize to experiment with a GPSDO simulator. For now I have the
>> following matlab function that "somewhat" does what I think is reasonable,
>> but I would like a reality check.
>> 
>> This is the matlab code:
>> 
>> function [phase] = synth_osc(samples,da,wn,fn)
>> # aging
>> phase = (((1:samples)/86400).^2)*da;
>> # white noise
>> phase += (rand(1,samples)-0.5)*wn;
>> # flicker noise
>> phase += cumsum(rand(1,samples)-0.5)*fn;
>> end
>> 
>> There are three components in the model, aging, white noise and flicker
>> noise, with everything expressed in fractions of seconds.
>> 
>> The first term basically creates a base vector that has a quadratic aging
>> function. It can be parametrized e.g. from an OCXO datasheet, daily aging
>> given in s/s per day.
>> 
>> The second term models white noise. It's just a random number scaled to the
>> desired 1-second uncertainty.
>> 
>> The third term is supposed to model flicker noise. It's basically a random
>> walk scaled to the desired magnitude.
>> 
>> As an example, the following function call would create a phase vector for a
>> 10MHz oscillator with one day worth of samples, with an aging of about 5
>> Millihertz per day, 10ps/s white noise and 10ns/s of flicker noise:
>> 
>> phase = osc_synth(86400, -44e-6, 10e-12, 10e-9);
>> 
>> What I'd like to know - is that a "reasonable" model or is it just too far
>> off of reality to be useful? What could be changed or improved?
>> 
>> Best regards,
>> Matthias
>> 
>> 
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[time-nuts] Re: Simple simulation model for an OCXO?

[Bob kb8tq]

Hi

….. except that having done this for many decades on hundreds of
designs, , a single data set from a real OCXO is likely to show you 
things that millions of simulations from a formula will somehow miss ….

Bob

> On May 2, 2022, at 5:13 PM, Greg Maxwell  wrote:
> 
> On Mon, May 2, 2022 at 10:01 PM Bob kb8tq  wrote:
>> By far the best approach is to use actual data. Grab a pair of OCXO’s and
>> compare them. A single mixer setup is one easy ( = cheap ) way to do it. You
>> will get the sum of the two devices, but for simulation purposes, it will be 
>> *much*
>> closer to reality than anything you can brew up with a formula.
> 
> But a programmatic way of generating plausible OCXO noise lets you
> generally millions of times more data, over a much larger spectrum of
> plausible operating conditions-- so that you can test the stability of
> an algorithm over a wider collection of conditions.
> 
> It's not a complete replacement for using real data-- you should do
> that too.  But it can be a much more comprehensive test.
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[time-nuts] Re: Simple simulation model for an OCXO?

[Bob kb8tq]

Hi

By far the best approach is to use actual data. Grab a pair of OCXO’s and 
compare them. A single mixer setup is one easy ( = cheap ) way to do it. You
will get the sum of the two devices, but for simulation purposes, it will be 
*much*
closer to reality than anything you can brew up with a formula. 

Bob

> On May 2, 2022, at 10:12 AM, Matthias Welwarsky  
> wrote:
> 
> Dear all,
> 
> I'm trying to come up with a reasonably simple model for an OCXO that I can 
> parametrize to experiment with a GPSDO simulator. For now I have the 
> following 
> matlab function that "somewhat" does what I think is reasonable, but I would 
> like a reality check.
> 
> This is the matlab code:
> 
> function [phase] = synth_osc(samples,da,wn,fn)
> # aging
> phase = (((1:samples)/86400).^2)*da;
> # white noise
> phase += (rand(1,samples)-0.5)*wn;
> # flicker noise
> phase += cumsum(rand(1,samples)-0.5)*fn;
> end
> 
> There are three components in the model, aging, white noise and flicker 
> noise, 
> with everything expressed in fractions of seconds.
> 
> The first term basically creates a base vector that has a quadratic aging 
> function. It can be parametrized e.g. from an OCXO datasheet, daily aging 
> given in s/s per day.
> 
> The second term models white noise. It's just a random number scaled to the 
> desired 1-second uncertainty.
> 
> The third term is supposed to model flicker noise. It's basically a random 
> walk scaled to the desired magnitude.
> 
> As an example, the following function call would create a phase vector for a 
> 10MHz oscillator with one day worth of samples, with an aging of about 5 
> Millihertz per day, 10ps/s white noise and 10ns/s of flicker noise:
> 
> phase = osc_synth(86400, -44e-6, 10e-12, 10e-9);
> 
> What I'd like to know - is that a "reasonable" model or is it just too far 
> off 
> of reality to be useful? What could be changed or improved?
> 
> Best regards,
> Matthias
> 
> 
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[time-nuts] Re: Can the ADEV of a GPSDO output ever be lower than the minimum of the ADEV of the internal oscilator and the ADEV of the GPS PPS?

[Bob kb8tq]

Hi

If you have a FS740 and measure it’s performance …. you likely will 
take anything the manual says a lot less seriously ….. Their ADEV 
performance in the real world is a bit underwhelming.

Bob

> On May 2, 2022, at 1:42 AM, Markus Kleinhenz via time-nuts 
>  wrote:
> 
> Hi Erik,
> 
> I just found a hint that what you are seeing may be correct after all:
> 
> The Manual of the Stanford Research Systems FS740 says:
> 
>/*Predictive Filtering*//
>//The superior short term stabilities of the OCXO and Rb timebases
>enable the usage of//
>//predictive filtering to improve the stability of the FS740 by up
>to 3 times over traditional//
>//methods. Predictive filtering uses state space methods to predict
>the phase of the local//
>//timebase relative to GNSS. The technique is quite similar to
>Kalman filtering. The//
>//benefit is that the FS740 can average the GNSS signal much more
>effectively, resulting//
>//in a significantly more stable signal with a much shorter time
>constant than would be//
>//possible with traditional filtering./
> 
> And has the ADEV Plots I attached. The GPS curve they printed is in the
> realm of a sawtooth corrected M8T (<1e-12 @ tau=10ks) [See the Plot from
> John Ackermanns Ublox evaluation]. But especially the Rb option seems to
> surpass the reference in a parallel fashion.
> 
> My two cents on simulated 1PPS Signals:
> 
> One has to be careful when only using ADEV as the only characteristic
> for modeling the 1PPS Signal as it combines White PM and Flicker PM in
> one slope. So you may create an artificial signal which is pure WPN and
> in turn is best predicted by something like the kalman filter.
> 
> Regards,
> 
> Markus
> 
> Am 29.04.2022 um 19:24 schrieb Erik Kaashoek:
>> Thanks for confirming something is still wrong. :-(
>> I've extended the simulation to contain a full Kalman filter working
>> with 2 state parameters: phase and frequency.
>> The biggest impact I can see is when increasing Kp above the optimal
>> value the PPS noise normally starts to impact the output phase and the
>> ADEV at tau 1 becomes worse
>> The Kalman filter seems to be able to filter the noise from the PPS
>> better so with equally high Kp the ADEV at tau =1 is about a factor 4
>> better
>> Unfortunately the high Kp of 0.1 is far from optimal and setting Kp to
>> 0.01 gives overall a better performance and the Kalman filter no
>> longer seem to have a visible impact.
>> Octave code for the simulation and the used data files are attached.
>> Also 3 plots are attached showing optimal Kp, high Kp with no filter
>> and high Kp with Kalman filer
>> I'm still seeing some weird stuff in the ADEV plots.
>> Erik.
>> 
>> On 29-4-2022 16:53, André Balsa wrote:
>>> Hi Erik,
>>> Mathematically, no, a GPSDO cannot have a lower uncertainty (ADEV)
>>> than the
>>> minimum observable uncertainty (ADEV) of the combined oscillator
>>> (disciplined clock) and PPS (disciplining clock) from the GPS receiver.
>>> Unless there is some magic trick to remove the uncertainty in a clock
>>> that
>>> I am not aware of. ;)
>>> 
>>> On Thu, Apr 28, 2022 at 10:03 PM Erik Kaashoek 
>>> wrote:
>>> 
 I'm doing some simulations to understand the impact of a filter
 between the
 TIC measurement and the PI controller steering the Vtune of the OCXO.
 With a well tuned PI controller without filter the best ADEV I can
 get is
 just above the minimum ADEV of an actual measured  OCXO and an actual
 measured GPS PPS.
 When I add an alpha-beta filter, similar to a first order Kalman filter
 with a manually tuned Kalman gain, and using similar Kp, Ki, the
 overall
 performance does not change (much)
 However with the filter its is possible to increase the Kp, Ki with a
 factor 10 and when I use in the simulation instead of a measured PPS an
 artificial PPS created from noise with the same ADEV as the GPS PP
 but with
 a very constant phase (different from the varying phase of a GPS
 PPS)  the
 ADEV of the GPSDO output in my simulation seems to drops below the
 ADEV of
 the PPS. Am I correct to assume this is a hint there is still something
 wrong in the simulation or was my initial assumption about the possible
 range of the GPSDO ADEV wrong?
 Erik.
 
> 
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[time-nuts] Re: What time difference to expect from two clocks using internal GPS receivers?

[Bob kb8tq]

Hi

If you are looking at time ( = the absolute offset from GPS’s version of UTC) 
then
there are a number of issues. 

The antenna you use will have a delay and it may well vary more than a bit. The 
cable to the antenna is in the same category. If both your modules run off a 
power 
splitter then those will not show up in an A-B comparison. 

The modules themselves likely have SAW filters in them. These have group delay 
just like any other bandpass filter. The tolerance on this is likely in the 
10’s of ns
module to module. There are other bit and pieces that can contribute at the 
“nanoseconds”
level. 

A typical set of modules from a good supplier should come in to a +/- 20 ns 
sort of
window for 2/3 of the parts ( = 1 sigma). 

Geometry errors are fairly simple. A meter is 3 ns in free space. Each meter 
you 
are off from “correct” will add 3 ns of “wobble” in the results. Just how this 
shows
up is very dependent on the direction of the error an what sort of satellite 
view you
happen to have. 

Ionosphere can ( in high sunspot years) contribute 50 ns or more to timing 
errors. 
Tropospheric issues can also get into the mix at a bit lower level. You might 
think
these would wash out on co-located units. Unfortunately they are not going to 
start /stop using this or that sat at exactly the same time. 

Lots of fun stuff to look for ….

Bob

> On Apr 30, 2022, at 6:41 AM, Erik Kaashoek  wrote:
> 
> Some more info
> The two GPS do keep their phase stable vs a Rb within +/-10 ns. But the
> absolute time difference of their PPS pulses  was, after a cold start,
> stable within +/- 20ns but  the average value could be up to 100ns and
> differed after every cold start.
> The two GPS antenna cables had a length difference of 1 meter, but that
> should cater for only 5 ns (?) One module is connected to the antenna with
> only a C, the other has a 1 GHz CLC high pass filter between antenna and
> module
> Erik
> 
> 
> 
> Op za 30 apr. 2022 om 12:32 schreef Erik Kaashoek :
> 
>> The PPS jitter of a cheap Chinese GPS module was measured at about +/-
>> 10 ns.
>> But the phase of the PPS compared to a Rb varied substantial more.
>> To verify if this was possibly due to ionospheric or atmospheric
>> conditions the time difference between the PPS of two identical modules
>> using two identical rooftop antenna was measured. Both only used the GPS
>> constellation.
>> This showed difference of up to 100 ns. Switching to GPS+GLN did not
>> make a visible difference.
>> It was tried to set both GPS modules into fixed position mode but the
>> reported position still kept moving a bit (within 3 m) and the fixed
>> mode did not have a visible impact on the time difference variations.
>> Is a time difference of up to 100 ns to be expected when using two GPS
>> receivers or is this difference possibly due to bad application or
>> performance of the cheap Chinese GPS modules
>> Erik.
>> 
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[time-nuts] Re: GPS Control Loop

[Bob kb8tq]

Hi

> On Apr 27, 2022, at 1:09 PM, ASSI  wrote:
> 
> On Dienstag, 26. April 2022 23:20:34 CEST Lux, Jim wrote:
>> That's the MCXO  - uses third harmonic and fundamental to measure the
>> temperature.  Q-Tech sells them. Or, more properly, has them in their
>> catalog and may be happy to quote a price and delivery. The datasheet
>> revisions are >5 years ago, except for the Space version which was
>> updated a couple years ago.  They're fairly good temp stability (a few
>> ppb over  -40 to +90) and lower power than a OCXO of comparable
>> performance.
> 
> Neat.  Well, they're certainly not on their stock list (nor the TCXO ultra-
> miniature that would drop into a rasPi).
> 
>> The fancy temperature compensation doesn't say anything
>> about aging, of course.
> 
> It seems that aging is a thing with this particular implementation, seeing 
> that they can use a reference to tune it out.  But they also seem to have 
> implemented a scheme that determines aging by taking the crystal out of the 
> synthesis loop for a little while.  The data-sheet dooesn't really say if 
> it's 
> only active at start-up nor whether the continuous correction requires a 
> reference.
> 
>> It's unclear whether cycling the temperature up
>> and down at room temp makes an oscillator age faster than one held at a
>> constant higher oven temp.
> 
> The control loop I implemented for my self-ovenized rasPi can do linear 
> sweeps 
> of the temperature (usually across the turnover point to determine or refine 
> the control parameters, you can set both the slope and the span).  I have no 
> hard data, but my impression is that both the sweeping itself and especially 
> the ramp speed contribute to an accelerated drift that takes a few days to 
> subside when returning to constant temperature.

As noted in a number of papers, the actual minimum temperature sensitivity
point on an OCXO will not be exactly at the turnover temperature. That will
be the crystal’s “best” but there are a number of other items in the OCXO that
get into the mix as the crystal sensitivity goes to zero. Modern OCXO production
uses temperature test data to set the OCXO to the “best” point. 

Bob

> 
> 
> Regarts,
> Achim.
> -- 
> +<[Q+ Matrix-12 WAVE#46+305 Neuron microQkb Andromeda XTk Blofeld]>+
> 
> SD adaptations for KORG EX-800 and Poly-800MkII V0.9:
> http://Synth.Stromeko.net/Downloads.html#KorgSDada
> 
> 
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[time-nuts] Re: GPS antenna locations

[Bob kb8tq]

Hi



> On Apr 17, 2022, at 12:58 PM, N1BUG  wrote:
> 
> Thanks Bob.
> 
> I have no idea what makes that gap to the south. There is nothing to block 
> that direction above 15 degrees.
> 
> I am not concerned about cable losses. I have LMR-400 on this roof mounted 
> antenna because it is only a 50 foot run, but the smallest cable I have to 
> either tower is 1/2". The antenna spec sheet says 30 dB gain.

The TBolt needs about 16 db in front of it to do a reasonable job. With a 
30 db gain antenna and no splitters, that leaves you with about 14 db or 
so for cable and connector loss. Most of us accumulate multiple GPS
gizmos. A 4 way splitter takes your 14 down to 8 db. That puts you “at
the limit” with a bit over 100’ of LMR-400.


> 
> The data sheet on this antenna shows it being down >60 dB at +/- 50 MHz. With 
> a noise figure spec of 2.2 dB I suspect that filter is before the amplifier, 
> but I wish it actually said that. Add several tens of dB for separation of 
> antennas and I think it would be OK.

The filter is after the amplifier. However the antenna itself is pretty 
narrowband. 
If the filter was in front, getting a 2 db noise figure would be exciting. 

Bob

> 
> My only concern is with overload of the onboard amplifier in the antenna. It 
> is filtered again by a GPS Networking splitter (-60 dB at +/- 60 MHz) before 
> going to the Thunderbolt.
> 
> Side mounted at 80 feet on the VHF tower (60 feet higher than where the one 
> on the roof is), it would have a much better sky view, neglecting any 
> blockage to the north from the tower itself and at high elevation angles from 
> yagis on top of the tower).
> 
> I am leaning toward getting another of these antennas and side mounting it on 
> the tower, then doing a comparison of the signal strength vs az/el plot 
> against this roof mounted one. It is an extra expense I will have to find a 
> way to budget for, but certainly has educational value if nothing else.
> 
> Paul
> 
> 
> 
> 
> 
> On 4/17/22 09:04, Bob kb8tq wrote:
>> Hi
>> Not knowing everything about the local environment there’s
>> not much way to guess exactly what this or that location
>> will do. The biggest thing I see on your plot is something
>> due south of the current antenna location.
>> In a “typical” setup, anything within 20 degrees of the horizon
>> gets tossed out for timing. The paths are long and with normal
>> clutter multi path is likely at low angles.
>> The filters in the typical “telecom” antennas are set up to block
>> cell phone transmitters. The GPS and cell antennas are co-located
>> on the same tower so they can get hit pretty hard. That said, the
>> cell site isn’t running an ERP in the many hundreds of watts range.
>> The longer your cable, the more likely you are to need a booster
>> amp. The telecom antennas typically don’t have a lot of gain. Yes,
>> fancy cable can help with this. RG-58 is a bad idea :) ….
>> If you have a better sky view at 60 to 80’ on the tower, then a side
>> mount in that range would be my vote. Lightning would be my
>> biggest concern as you go higher. Assuming all the heights are
>> to the same reference, moving the antenna up 40 to 60’ should
>> do the trick.
>> Bob
>>> On Apr 16, 2022, at 2:05 PM, N1BUG  wrote:
>>> 
>>> Hello time nuts,
>>> 
>>> I finally have my long awaited Trimble Thunderbolt up and running. I am not 
>>> thrilled with the coverage using a Symmetricom 58532A antenna on the roof 
>>> about 20 feet above ground level. Here is what I get after ~24 hours:
>>> 
>>> http://n1bug.com/gpssig.png
>>> 
>>> Sometimes I have 8 satellites with usable signal, sometimes as few as 5. 
>>> The problem to the west is trees. I believe the chaotic signal strength in 
>>> the east is due to reflections from a metal roof.
>>> 
>>> I have three options:
>>> 
>>> 1. Leave the antenna where it is.
>>> 
>>> 2. Side mount it at 80 to 90 feet on a radio tower that has yagis for 
>>> 50/432/222/144 MHz at 105/110/115/120 feet. These antennas are used for 
>>> high power transmitting. Potential interference to GPS reception? I don't 
>>> know if the filter in the 58532A is before or after the amplifier. Blockage 
>>> from the tower and/or yagis? I assume mounting a few feet off the south 
>>> tower face would be best.
>>> 
>>> 3. Mount at the top of a mast on another radio tower, at 110 feet. This 
>>> would have a completely unobstructed sky view but would have antennas for 
>>> 7/10 MHz about 3 feet below and 14/18/21/24/28 MHz about 13 

[time-nuts] Re: GPS antenna locations

[Bob kb8tq]

Hi

Not knowing everything about the local environment there’s 
not much way to guess exactly what this or that location 
will do. The biggest thing I see on your plot is something 
due south of the current antenna location. 

In a “typical” setup, anything within 20 degrees of the horizon
gets tossed out for timing. The paths are long and with normal
clutter multi path is likely at low angles. 

The filters in the typical “telecom” antennas are set up to block
cell phone transmitters. The GPS and cell antennas are co-located
on the same tower so they can get hit pretty hard. That said, the
cell site isn’t running an ERP in the many hundreds of watts range. 

The longer your cable, the more likely you are to need a booster
amp. The telecom antennas typically don’t have a lot of gain. Yes,
fancy cable can help with this. RG-58 is a bad idea :) ….

If you have a better sky view at 60 to 80’ on the tower, then a side
mount in that range would be my vote. Lightning would be my 
biggest concern as you go higher. Assuming all the heights are 
to the same reference, moving the antenna up 40 to 60’ should 
do the trick. 

Bob

> On Apr 16, 2022, at 2:05 PM, N1BUG  wrote:
> 
> Hello time nuts,
> 
> I finally have my long awaited Trimble Thunderbolt up and running. I am not 
> thrilled with the coverage using a Symmetricom 58532A antenna on the roof 
> about 20 feet above ground level. Here is what I get after ~24 hours:
> 
> http://n1bug.com/gpssig.png
> 
> Sometimes I have 8 satellites with usable signal, sometimes as few as 5. The 
> problem to the west is trees. I believe the chaotic signal strength in the 
> east is due to reflections from a metal roof.
> 
> I have three options:
> 
> 1. Leave the antenna where it is.
> 
> 2. Side mount it at 80 to 90 feet on a radio tower that has yagis for 
> 50/432/222/144 MHz at 105/110/115/120 feet. These antennas are used for high 
> power transmitting. Potential interference to GPS reception? I don't know if 
> the filter in the 58532A is before or after the amplifier. Blockage from the 
> tower and/or yagis? I assume mounting a few feet off the south tower face 
> would be best.
> 
> 3. Mount at the top of a mast on another radio tower, at 110 feet. This would 
> have a completely unobstructed sky view but would have antennas for 7/10 MHz 
> about 3 feet below and 14/18/21/24/28 MHz about 13 feet below. Those antennas 
> are used for high power transmitting. There will at some point be a 10 GHz 
> dish about 8 feet below the top of that mast.
> 
> Any comments on these options? Is it good enough where it is? I am only using 
> it as a 10 MHz reference now, but I may care about the 1 PPS later.
> 
> Paul N1BUG
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[time-nuts] Re: Question about GPS 1PPS ADEV

[Bob kb8tq]

Hi

Any time you look inside a control loop, there will be bumps and 
the like that are a function of the control loop plus the dynamics
of the device being controlled plus the environment. If you have
a straight line … something is wrong. 

Bob

> On Apr 16, 2022, at 1:20 PM, Matthias Welwarsky  
> wrote:
> 
> On Samstag, 16. April 2022 17:55:17 CEST John Ackermann N8UR wrote:
>> Finally, it looks like you're comparing the raw PPS with an oscillator 
>> that is steered by that same PPS.  I have to think their correlation 
>> could lead to possible and unpredictable errors.  It would be better to 
>> have the OCXO remain unsteered during the measurement.
> 
> That's correct. But I'm not trying to characterize the OCXO or the GNSS 
> receiver. I was just wondering about this perceived discrepancy.
> 
>> 
>> Best,
>> John
>> 
>> 
>> 
>>> Best regards,
>>> Matthias
>>> 
>>> https://hamsci.org/sites/default/files/publications/2020_TAPR_DCC/
>>> N8UR_GPS_Evaluation_August2020.pdf
>>> 
>>> 
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[time-nuts] Re: Question about GPS 1PPS ADEV

[Bob kb8tq]

Hi

> On Apr 16, 2022, at 11:55 AM, John Ackermann N8UR  wrote:
> 
> 
> 
> On 4/16/22 09:52, Matthias Welwarsky wrote:
>> Dear list members,
>> in 2020 John Ackermann published an evaluative survey of current day GPS and
>> GNSS receivers (URL below). I have a question about figure 26, which shows,
>> among others, the ADEV of a NEO-M8T against a Cesium reference, with
>> quantization correction applied. The curve shows a significant "bulge" above
>> 1e-10 between about 10s and 100s tau.
>> I'm using a LEA-M8T in my DIY GPSDO, which I think is the same chipset in a
>> slightly different package. I have attached an image worth about 3000 seconds
>> of data, raw 1PPS phase difference against the LO used in the GPSDO. The 
>> GPSDO
>> is locked, not in hold-over mode.
>> The bulge between 10s and 100s is not really visible here. There is a slight
>> bend, but not as pronounced. My explanation is that this is due to the LO
>> being pulled by the GNSS receiver so that it is no longer fully visible. I
>> reason that, were the LO more stable, more loosely coupled to the GNSS, I
>> should see the bulge from figure 26. Would you agree?
> 
> My theory of the ADEV flat spot when the M8T is qErr corrected is that 
> multiple things contribute to noise on the PPS output, and the qErr is only 
> one of them, and it consists of a noise source (clock granularity) that is 
> unrelated to any external analog process.
> 
> Since Qerr has a fixed limit of ± XX nanoseconds (half the receiver clock 
> granularity), its contribution to the PPS noise decreases with longer 
> averaging times.  Meanwhile, the other sources of noise such as ionosphere, 
> etc., are slower and become more pronounced as tau increases.  At around 30 
> seconds, the external noise factors become larger than the qErr.
> 
> So at short tau cancelling out the qErr gets rid of a major noise source and 
> improves ADEV but as tau increases beyond ~30 seconds the qErr contribution 
> is gradually outweighed by the other noise sources and disappears, returning 
> the ADEV slope to its normal -1.
> 
> Also note that in Fig. 26, at tau below 3 seconds it's possible that both the 
> M8T and F9T corrected plots are limited by the TICC resolution, which is 
> around 8e-11 @ 1 second on a good day.  That's below these traces, but may 
> still be high enough to impact the measurement.
> 
> Finally, it looks like you're comparing the raw PPS with an oscillator that 
> is steered by that same PPS.  I have to think their correlation could lead to 
> possible and unpredictable errors.  It would be better to have the OCXO 
> remain unsteered during the measurement.

The only way to make meaningful single comparison measurements of what’s 
going on is to compare to an external “free running” standard that has a 
stability 
adequate to the task. Normally this means an ADEV 5 to 10X better than what
you expect to see on the device under test. 

Bob

> 
> Best,
> John
> 
> 
>> Best regards,
>> Matthias
>> https://hamsci.org/sites/default/files/publications/2020_TAPR_DCC/
>> N8UR_GPS_Evaluation_August2020.pdf
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[time-nuts] Re: General discussion of PID algorithms applied to GPSDO control loops (continued 1)

[Bob kb8tq]

Hi

Even if you *do* have a number for your OCXO EFC, what is that number?

The nominal sensitivity is fine. The next layer is the ratio of maximum slope
to minimum slope. This is a very different thing than the “linearity” that is 
normally
specified. Even the linearity is something you need to look carefully at the 
Mil-O-55310 definition. It’s not quite what you would expect it to be.

Bottom line: the sensitivity can easily vary 2:1 or even 4:1 on a pretty good 
part. 

Bob

> On Apr 15, 2022, at 5:13 PM, Skip Withrow  wrote:
> 
> One of the things that cannot be ignored are the statements made in
> item #10 regarding the DAC and OCXO EFC.  If the DAC is 16 bits that
> is 2^16=65,536 steps.  If the DAC outputs 0-4 volts that is 61uV/step.
> 
> Now you have to take this number and multiply it by your OCXO
> sensitivity (not given), and you have the minimum frequency step that
> can be made.
> 
> This is the best that you can ever do.  So, your TIC/1pps resolution
> does not need to be any better than this.  Waiting a long time to get
> higher resolution may (probably) buys you nothing.
> 
> This is where all the magic comes in - higher resolution DACs, more
> stable oscillators, more stable GPS PPS, etc.
> 
> To a first order, overlaying the line from item #6 with the ADEV of
> the oscillator used and setting the loop time constant at the
> crossover tau works pretty good.
> 
> As has been said many times, if it was easy everybody would be doing it.
> 
> Regards,
> Skip Withrow
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[time-nuts] Re: General discussion of PID algorithms applied to GPSDO control loops (continued 1)

[Bob kb8tq]

Hi


> On Apr 15, 2022, at 8:36 AM, André Balsa  wrote:
> 
> Hi Bob, thank you for your comments, very helpful.
> 
> On Fri, Apr 15, 2022 at 2:06 PM Bob kb8tq  wrote:
> 
>> Hi
>> 
>>> For example, the STM32 GPSDO with
>>> its very crude FLL algorithm used a fixed interval of 429s to change Vctl
>>> (except during the initial calibration). In older GPSDO designs using a
>>> purely analog circuit the control variable is applied continuously.
>> 
>> At 429 s, you would need to have a *very* low drift oscillator or put up
>> with a lot of noise on the output.
> 
> 
> Since the STM32 GPSDO uses an OCXO it does indeed have a very low drift
> oscillator.

Relative to a 429 second measurement period, a maser would be a low drift
device. A typical OCXO is very high drift at this sort of tau.

> 
>> 
>> 
>>> 
>>> 12. Consequently the program of an FLL or PLL loop for a GPSDO has two
>>> decisions to make every second: a) what is the size of the correction to
>> be
>>> applied to the control variable iow what is the new value for Vctl ? and
>> b)
>>> Should the correction that was just computed be applied now, or should we
>>> wait and apply a different correction later ?
>> 
>> If the loop only has useful information every 429 seconds, then there
>> is not much value in updating any more often. This is one thing that makes
>> PLL’s the more common approach.
>> 
> Actually the FLL loop has useful information every second, so we can update
> Vctl at any time interval that is a multiple of 1s, or even at varying time
> intervals. That is exactly the question that I am asking.

If the data all is “back 429 seconds” then (practically speaking ) you only 
have 
one data point per 429 seconds. 

> 
> 
>>> 
>>> 13. The programmer (in this case myself) has to decide whether to use a
>> P,
>>> PI or PID loop, the optimal values for Kp, Ki and Kd, the use of a fixed
>> or
>>> variable "time constant" (the delay between changes to the Vctl), and any
>>> processing (filetring, averaging, removal of outlying values, etc) of the
>>> measurements from the frequency counter or the TIC.
>> 
>> The parameters will be highly dependent on exactly what you have in your
>> setup. Large amounts of damping is normally good. A crossover frequency
>> for the system that makes sense vs the measured noise is generally the
>> next thing on the list.
> 
> 
> Indeed, I guess any algorithm I can come up with will have to pay close
> attention to the inherent noise/jitter in the readings. Thanks for pointing
> this out.

The only way to know the noise profiles is to measure them against an
external “better than” standard. There is no practical way to come up with
the info from “inside” the device. 

> 
>> 
>> 
>>> 
>>> 14. More precisely in my case, there is an extra complicating factor
>>> because I am trying to merge the FLL and PLL control loops into a
>> "hybrid"
>>> FLL/PLL control loop. How to make the best use of the information from
>> the
>>> two measurement ?
>> 
>> The loop needs to be either an FLL or a PLL. It is not at all unusual to
>> switch
>> out the entire loop as the GPSDO “warms up”.
>> 
>> Bob
>> 
>> You mean the FLL and PLL are exclusive of each other ? I guess you are
> right, but I am trying to think "outside the box" and see if there are any
> alternatives.

You will have two people driving the car at the same time. One hits the 
accelerator and the other hits the brakes at the same time. They both can’t
be active *and* feed the EFC at the same time. The practical answer is to
run each during the warmup phase that it makes sense to do so.

Bob

> 
> Again, thanks a bunch for your comments.
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[time-nuts] Re: General discussion of PID algorithms applied to GPSDO control loops (continued 1)

[Bob kb8tq]

Hi

> On Apr 14, 2022, at 12:00 PM, André Balsa  wrote:
> 
> 10. Again referring to our ideal PID controller, our control variable u(t)
> in the case of most low-cost GPSDOs is the control voltage that we can
> apply to the OCXO to vary its frequency. For Lars' DIY GPSDO and for the
> STM32 GPSDO a 16-bit PWM DAC is used to generate a control voltage Vctl
> between 0 and 4V. This is an extremely economical solution but it does the
> job.
> 
> 11. A complicating factor for programming the PLL or FLL algorithms for a
> GPSDO is that we are dealing with a discrete correction process, in other
> words changes in the control variable are applied at discrete time
> intervals which can be fixed or variable. For example, the STM32 GPSDO with
> its very crude FLL algorithm used a fixed interval of 429s to change Vctl
> (except during the initial calibration). In older GPSDO designs using a
> purely analog circuit the control variable is applied continuously.

At 429 s, you would need to have a *very* low drift oscillator or put up
with a lot of noise on the output. 

> 
> 12. Consequently the program of an FLL or PLL loop for a GPSDO has two
> decisions to make every second: a) what is the size of the correction to be
> applied to the control variable iow what is the new value for Vctl ? and b)
> Should the correction that was just computed be applied now, or should we
> wait and apply a different correction later ?

If the loop only has useful information every 429 seconds, then there 
is not much value in updating any more often. This is one thing that makes
PLL’s the more common approach.

> 
> 13. The programmer (in this case myself) has to decide whether to use a P,
> PI or PID loop, the optimal values for Kp, Ki and Kd, the use of a fixed or
> variable "time constant" (the delay between changes to the Vctl), and any
> processing (filetring, averaging, removal of outlying values, etc) of the
> measurements from the frequency counter or the TIC.

The parameters will be highly dependent on exactly what you have in your
setup. Large amounts of damping is normally good. A crossover frequency
for the system that makes sense vs the measured noise is generally the 
next thing on the list. 

> 
> 14. More precisely in my case, there is an extra complicating factor
> because I am trying to merge the FLL and PLL control loops into a "hybrid"
> FLL/PLL control loop. How to make the best use of the information from the
> two measurement ?

The loop needs to be either an FLL or a PLL. It is not at all unusual to switch
out the entire loop as the GPSDO “warms up”.

Bob

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[time-nuts] Re: +1/f of transistors

[Bob kb8tq]

Hi

As noted in many posts, 1/F noise is dependent on a number of
operating parameters. The first thing to do would be to understand
your use case. Testing under one set of conditions may not be 
useful for use in other conditions.

Bob

> On Apr 10, 2022, at 5:50 AM, Leon Pavlovic  wrote:
> 
> What would be a (very) good lab setup for the 1/f noise evaluation? I'm
> talking about BJTs and JFETs.
> 
> What instrumentation and power supply are needed? How not to measure in a
> wrong way - like 1/f noise in the instrumentation or the power supply
> circuitry? Active collector/drain current sources vs pure resistive...
> 
> I'm quite familiar with the low NF measurements, so I guess the same
> practice would apply: insert your DUT, short its input side, have a
> low-noise post amp and display the noise on FFT SA?
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[time-nuts] Re: Low Phase Noise 10 MHz bench signal source sought

[Bob kb8tq]

Hi

Crystal filtering for far removed phase noise does work. It’s far from a 
“perfect”
solution. The crystal it’s self contributes noise and it does AM to PM 
conversion. 
You pretty much always get a “noise hump” as a result. 

Like any filter, a single crystal filter has an attenuation vs frequency curve. 
You
can sweep it with a networks analyzer to see this. A simple series crystal will 
be
a very “lopsided” filter. There are well known added bits and pieces that can 
impact this. 

Any filter will have an input/output impedance vs frequency as well as an 
attenuation 
vs frequency characteristic. This change likely will have an impact on the noise
of whatever amplifier that follows the filter. 

Coming up with “sub thermal” noise this way assumes that the post amplifier 
has some pretty unusual characteristics. As noted in a lot of places, the -174 
dbc/Hz
we typically play with assumes a 50 ohm system. If you have a 1 ohm system …
thats’s a different thing.

Bob



> On Apr 8, 2022, at 8:06 AM, Mike Monett  wrote:
> 
> Dear Fans,
> 
> While I was researching cross-correlation, I came across an interesting
> paper that described different methods of reducing OCXO phase noise. They
> involved putting a 10 MHz crystal between the buffer amplifier and the
> output load. One method promised "sub-thermal" phase noise. The idea is the
> crystal impedance rose sharply off resonance, so the energy at those
> frequencies was heavily attenuated.
> 
> I don't know how much additional noise the crystal ESR would introduce, but
> it is an intriguing idea and might be worth pursuing.
> 
> MRM
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[time-nuts] Re: Low Phase Noise 10 MHz bench signal source sought

[Bob kb8tq]

Hi

Since this is Time Nuts … crazy isn’t always ruled out :)

> ……….
> “TSP #162   Tutorial   on   Theory,   Characterization  & Measurement
> Techniques of Phase Noise"
> 
>  https://www.youtube.com/watch?v=SOHjFtw0sgo
> 
> each 5db  of improvement requires an order of magnitude  increase in
> the number of correlations.
> 
> dB  5   10  15  20  25  30  35  40
> N   10  100 10001E4 1E5 1E6 1e7 1e8
> 
> So going from -180dBc/Hz to 220dBc/Hz would require 1e8 correlations.
> 
> 6. Nobody would wait that long. But how many correlations do you need?
> ……..

Say you did go the brute force approach. We’re talking far removed
phase noise for stuff like -220 dbc/Hz so each sample set could be pretty 
short. 
Just to toss out a number, let’s go with 1 ms. Yes, this would be quite far 
from carrier.

We now are at 1x10^5 seconds of data. That’s “only” a bit over a day. That’s 
not 
an insane run time for an R environment. An “over the weekend” run of 
three days would get you all the way out to 3 ms sample sets. 

Want longer samples? 

How many samplers / correlation processors do you have? How much 
does each one add to the mix? If it was linear, you are out to a whopping
300 ms with “only” 100 processors. Yes that sounds insane. This isn’t quite
as simple as a video graphics card or four being plugged into a PC. Still, 
it could cut down the time. 

So maybe not quite so crazy after all …..

Bob
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[time-nuts] Re: GPScon peogram

[Bob kb8tq]

Hi

Since some of these programs are indeed a bit tough to find
(and when found run on this or that machine), It might be 
useful to know what the target device is. There are a lot of
programs out there and a good deal of overlap in terms of
what they support. 

Bob

> On Apr 5, 2022, at 7:33 PM, Lon Cottingham  wrote:
> 
>  How does one acquire GPScon?  The only source I find in from Jackson's 
> Labs ant it only works with Jackson Lab's products. 73 de Lon, K5JV 
> [c59de0c0-bb5f-4d7f-8c82-58c878259afb]
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[time-nuts] Re: Low Phase Noise 10 MHz bench signal source sought

[Bob kb8tq]

HI

(see below)

> On Apr 5, 2022, at 6:50 PM, Joseph Gwinn  wrote:
> 
> time-nuts Digest, Vol 216, Issue 5
> 
> Multiple responses interspersed below.
> 
>> --
>> 
>> Message: 1
>> Date: Sun, 3 Apr 2022 10:13:44 +0200
>> From: "Bernd Neubig" 
>> Subject: [time-nuts] Re: Low Phase Noise 10 MHz bench signal source
>>  sought
>> To: "'Discussion of precise time and frequency measurement'"
>>  
>> Message-ID: <003a01d84732$bcc47c40$364d74c0$@t-online.de>
>> Content-Type: text/plain;   charset="utf-8"
>> 
>> Hi,
>> 
>> This is nearly "off-the shelf" model:
>> https://www.axtal.com/cms/docs/doc100916.pdf
>> The ULN version does meet the -170 dBc/Hz
> 
> This unit is certainly plausible. 
> 
> 
>> --
>> 
>> Message: 4
>> Date: Sun, 3 Apr 2022 09:53:18 -0400
>> From: Bob kb8tq 
>> Subject: [time-nuts] Re: Low Phase Noise 10 MHz bench signal source
>>  sought
>> To: ew , Discussion of precise time and frequency
>>  measurement 
>> Message-ID: <11376923-062a-4011-a6d4-1d9ce3361...@n1k.org>
>> Content-Type: text/plain;   charset=utf-8
>> 
>> Hi
>> 
>> These days a PLL is going to either be analog or digital. If it’s 
>> analog, you get into size constraints related to capacitors
>> as you go to lower crossover frequencies. With digital, you
>> get into all of the noise issues that any digital circuit will have.
>> (Yes, they can be addressed but it’s not easy at very low
>> offset frequencies). 
> 
> All of the loop filters I've seen recently had nominal bandwidths in 
> the Hertz 
> to tens of Hertz, usually implemented in some kind of digital signal 
> processor.  

10 Hz or higher is certainly do-able with analog loop components. 
There are a lot of products out there that work that way.

> 
> About 30 years ago, there was a legacy 5 MHz disciplined 
> oscillator that could be set to a 100-second response time.  I never 
> did find any real technical data or patents on it.  I don't recall 
> its name, but it may come back to me.  I think it was made by 
> Symmetricom.

If size is not an issue, you can go to some pretty large value 
(likely Teflon) capacitors for an analog loop. Sourcing those 
parts today is likely to both be difficult and quite expensive. 

> 
> 
>> Regardless of design, you will always have some noise peaking.
>>> 89 degree phase margins can help with this, but they bring
>> in other problems. Setting the phase margin and other parameters
>> is either a mathematical design process or done with simulation. 
>> It can be very frustrating doing it by trial and error. I still find 
>> Phaselock Techniques by Floyd M Gardner to be a good reference 
>> on this stuff. 
>> 
>> The normal PLL control loops are fairly low order filters. If you
>> go high order, loop stability (and peaking) becomes difficult
>> to handle. Because it’s a simple filter, things like close in spurs 
>> will only be attenuated by some finite amount. 
> 
> I always suspected that that legacy disciplined oscillator used a 
> 3rd-order PLL loop, because it became unstable if the incoming 
> reference signal was too faint.  But that filter had to be their 
> secret sauce.

In a normal PLL, the input signals are pretty well defined in terms
of level. If you have some sort of external reference setup, then 
it is not at all uncommon to drop a limiter into the mix. 

Bob

> 
> 
> Joe Gwinn
> 
> 
> 
>> So yes, you will have issues. Dealing with those issues means 
>> an area ( = range of offset frequencies) that are not as nice as you
>> might wish. That’s just the way the real world works. 
>> 
>> Bob
>> 
>>> On Apr 3, 2022, at 8:14 AM, ew via time-nuts 
>>>  wrote:
>>> 
>>> There is a lot of talk of Rb's and OCXO's  and using an OCXO for 
>>> clean up. Very little about the clean up loop. It is key for 
>>> overall performance. We are working on it for quite some time and 
>>> are not happy with the results. On list off list would greatly be 
>>> appreciated.  
>>> Bert Kehren  
>> 
>> --
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[time-nuts] Re: 32.768Khz Crystal Trimming

[Bob kb8tq]

Hi

The cap ratio very much depends on how the internals of the chip
got done. Since you have zero info on that, it’s best to stick with the
ratios and approximate values shown in the app notes. 

The most common way to get a fine grain tune is to put something 
large in series with the crystal (maybe 120 or 240 pf)  and then pad 
it with small values (maybe 0.5 to 10 pf). Yes, we just took up more 
board space and upped the BOM … sorry about that :)

Bob

> On Apr 4, 2022, at 2:03 PM, Dan Kemppainen  wrote:
> 
> After playing with cap values the crystal is running about 6.5ppm fast. 
> Plenty good for what it is. Of course this was with a sample size of 1.
> 
> Being a time-nut, I did try a few different cap values just to see if it 
> could be made any better. The Next closest cap combo gives 8.3ppm slow.
> 
> Putting a little heat on the crystal showed it tended to drop in frequency, 
> so 6.5ppm fast is probably better than 8.3ppm slow.
> 
> Looking at the slope/intercept of the values tried, the needed value isn't 
> easily achievable. Again, n=1, so sample to sample variation is untested.
> 
> It looks like the 'ideal' cap value might be achievable, but that would 
> require a pretty big mismatch between C1/C2. More than 2:1, probably closer 
> to 3:1. I'm guessing that would be a 'no-no'?
> 
> Anyway, Thanks to all who replied!
> 
> Dan
> 
> 
> 
> 
> 
> 
> On 4/2/2022 3:27 AM, time-nuts-requ...@lists.febo.com wrote:
>> All,
>> Ahhh, Yes. Checking notes, during the initial build I slapped a pair of 10pF 
>> caps down (all I had on hand) just to get the RTCC oscillators somewhat 
>> working. I can see now that is WAY under needed capacitance.
>> Measurements are taken right in the micro, so no external leads to influence 
>> the board. A serial debug line spits out the difference in elapsed time for 
>> the RTCC and time derived from the 10Mhz EXT REF (GPSDO). (No extra 
>> wires/leads etc.)
>> Honestly this RTCC doesn't need to be stellar. It's more of a holdover 
>> oscillator when the EXT CLK is lost. But, this is time-nuts and all, so 
>> might as well squeeze everything out of it! :)
>> Bernd,
>> Your post, and rule of thumb are what I was looking for. I will certainly 
>> archive this in my notes. This should be good enough to save destroying the 
>> board by soldering/de-soldering many 0603 SMT caps.
>> Thanks to all who replied. This should be enough to keep me busy over the 
>> weekend!
>> Dan
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[time-nuts] Re: Commercial solution - 122.88 Mhz low noise source

[Bob kb8tq]

Hi

That’s a massive step down from what’s in there now. How much 
that impacts you depends a lot on what the end use is.  The plots
shown are in the vicinity of 10 MHz. If a straight multiply rule applies,
the performance at 122 MHz would be about 20 db worse. The chip 
datasheet:

https://www.skyworksinc.com/-/media/SkyWorks/SL/documents/public/data-sheets/Si5328.pdf
 
<https://www.skyworksinc.com/-/media/SkyWorks/SL/documents/public/data-sheets/Si5328.pdf>

suggests that this is likely to be true for offsets < 100 KHz.


No, this isn’t in any way a knock on that GPSDO. It’s doing what it
was designed to do. I’m sure it works very well. It simply isn’t 
targeted at the same sort of noise level as what’s on there now. 

Bob

> On Apr 4, 2022, at 11:35 AM, Jeff Blaine  wrote:
> 
> I found a commercial programmable GPSDO that looks like it would do the job.  
> Based on a Si chip.
> 
> http://leobodnar.com/files/Informal%20Evaluation%20of%20a%20Leo%20Bodnar%20GPS%20Frequency%20Reference.pdf
> 
> Performance is not in the "great" category but I am guessing it would be in 
> the "good enough" category.  And it does not require any design work or 
> project construction time.
> 
> 73/jeff/ac0c
> alpha-charlie-zero-charlie
> www.ac0c.com
> 
> 
> On 4/4/22 7:36 AM, Bob kb8tq wrote:
>> Hi
>> 
>> For a variety of reasons, you will have a loop bandwidth in the > 10 Hz
>> to < 200 Hz region. The VCXO isn’t great close in and the GPSDO (even
>> after multiplication) should be as good / better. Even with some work,
>> the VCXO still will wander around due to temperature. Unless you go
>> crazy, it also will wander a bit from supply. There’s also the chance of
>> vibration or shock.
>> 
>> As mentioned in another post, component values on an properly designed
>> analog loop running one of the typical chips get pretty crazy at very low
>> loop bandwidths. You now get into questions about sensitivity of components
>> if you go with certain ceramics, leakage if you run an electrolytic,  and 
>> with
>> size if you go with something else.
>> 
>> Bandwidth and damping ( or phase margin ) are two independent variables
>> when you design a PLL. You can have a narrowband bandwidth and a lot
>> of damping or the same bandwidth without much damping. It’s all in the
>> design equations ….
>> 
>> Bob
>> 
>>> On Apr 3, 2022, at 11:22 PM, Jeff Blaine  wrote:
>>> 
>>> It looks like the XO version is mounted on the PCB.  But there is a VCXO 
>>> version which would run about +/- 35 PPM/3.3V typical and that's 
>>> (amazingly) available.
>>> 
>>> Since this EFT is not an on-board capability, at the moment, I'm thinking 
>>> to mount the VCXO - and associated goodies to slave it to the 10 Mhz GPSDO 
>>> feed - in a separate enclosure that could allow the VCXO to be thermally 
>>> insulated.  The temp swing vs. time will move very slowly and that would 
>>> allow a very damped PLL loop to apply corrections.
>>> 
>>> I've not done anything with a PLL before but I don't envision this as being 
>>> too complicated.  And it would be a pretty cool project.  Don't need true 
>>> "phase" lock, but rather "frequency control" without degrading the noise 
>>> characteristics of the VCXO.
>>> 
>>> 73/jeff/ac0c
>>> alpha-charlie-zero-charlie
>>> www.ac0c.com
>>> 
>>> 
>>> On 4/3/22 7:38 PM, Lux, Jim wrote:
>>>> On 4/3/22 5:09 PM, Bob kb8tq wrote:
>>>>> Hi
>>>>> 
>>>>> Does the Abricon have an EFC input? If so, rigging up a PLL to lock
>>>>> 122.88 to 10 MHz is probably the best option.
>>>>> 
>>>>> Bob
>>>> 
>>>> And what about moving off the board - so you don't have to chase the 
>>>> temperature swings.
>>>> 
>>>> 
>>>>>> On Apr 3, 2022, at 7:57 PM, Jeff Blaine  wrote:
>>>>>> 
>>>>>> I've got a couple of Red Pitaya 122-16 SDR and want to discipline them 
>>>>>> to an external low noise GPSDO source.  Wanted to see if there were some 
>>>>>> easy solutions that fell into this category.
>>>>>> 
>>>>>> For the prior generation (QS1R), I had built a homebrew OCXO based on a 
>>>>>> custom low noise crystal from ICM and that worked great for the 125 Mhz 
>>>>>> application.  Unfortunately this Pitaya is a bit removed in Fc and ICM 
>>>>>> has been out of 

[time-nuts] Re: Commercial solution - 122.88 Mhz low noise source

[Bob kb8tq]

Hi

For a variety of reasons, you will have a loop bandwidth in the > 10 Hz 
to < 200 Hz region. The VCXO isn’t great close in and the GPSDO (even
after multiplication) should be as good / better. Even with some work,
the VCXO still will wander around due to temperature. Unless you go 
crazy, it also will wander a bit from supply. There’s also the chance of
vibration or shock. 

As mentioned in another post, component values on an properly designed
analog loop running one of the typical chips get pretty crazy at very low
loop bandwidths. You now get into questions about sensitivity of components
if you go with certain ceramics, leakage if you run an electrolytic,  and with 
size if you go with something else.

Bandwidth and damping ( or phase margin ) are two independent variables
when you design a PLL. You can have a narrowband bandwidth and a lot
of damping or the same bandwidth without much damping. It’s all in the 
design equations …. 

Bob

> On Apr 3, 2022, at 11:22 PM, Jeff Blaine  wrote:
> 
> It looks like the XO version is mounted on the PCB.  But there is a VCXO 
> version which would run about +/- 35 PPM/3.3V typical and that's (amazingly) 
> available.
> 
> Since this EFT is not an on-board capability, at the moment, I'm thinking to 
> mount the VCXO - and associated goodies to slave it to the 10 Mhz GPSDO feed 
> - in a separate enclosure that could allow the VCXO to be thermally 
> insulated.  The temp swing vs. time will move very slowly and that would 
> allow a very damped PLL loop to apply corrections.
> 
> I've not done anything with a PLL before but I don't envision this as being 
> too complicated.  And it would be a pretty cool project.  Don't need true 
> "phase" lock, but rather "frequency control" without degrading the noise 
> characteristics of the VCXO.
> 
> 73/jeff/ac0c
> alpha-charlie-zero-charlie
> www.ac0c.com
> 
> 
> On 4/3/22 7:38 PM, Lux, Jim wrote:
>> On 4/3/22 5:09 PM, Bob kb8tq wrote:
>>> Hi
>>> 
>>> Does the Abricon have an EFC input? If so, rigging up a PLL to lock
>>> 122.88 to 10 MHz is probably the best option.
>>> 
>>> Bob
>> 
>> 
>> And what about moving off the board - so you don't have to chase the 
>> temperature swings.
>> 
>> 
>>> 
>>>> On Apr 3, 2022, at 7:57 PM, Jeff Blaine  wrote:
>>>> 
>>>> I've got a couple of Red Pitaya 122-16 SDR and want to discipline them to 
>>>> an external low noise GPSDO source.  Wanted to see if there were some easy 
>>>> solutions that fell into this category.
>>>> 
>>>> For the prior generation (QS1R), I had built a homebrew OCXO based on a 
>>>> custom low noise crystal from ICM and that worked great for the 125 Mhz 
>>>> application.  Unfortunately this Pitaya is a bit removed in Fc and ICM has 
>>>> been out of biz for several years now.
>>>> 
>>>> The RP's on-board oscillator is the respectable Abracon ABLNO 122.88 which 
>>>> runs about -115 dBc/hz at 100 Hz spacing and that's probably not a 
>>>> limitation for the unit.  Unfortunately the OSC is a bit too sensitive to 
>>>> external temp and wanders around quite a lot (+/- 5 PPM) compared to the 
>>>> homebrew OCXO - over the annual ambient temp range of (10-35C) (all 
>>>> datasheet referenced values). The Abracon unit is mounted on the board 
>>>> with the FPGA and sees additional temp range depending on other heat 
>>>> generation factors (FPGA loading).
>>>> 
>>>> I have a couple of commercial GPSDO on the bench now, so a pretty clean 10 
>>>> Mhz GPSDO reference is available now.  And I've seen enough board 
>>>> discussion to know that a homebrew GPSDO solution is one of those "it's 
>>>> harder than it looks" things.  Hence the desire just to buy whatever is 
>>>> the silver bullet, as long as it does not require too much silver.  ha ha
>>>> 
>>>> Appreciate any suggestions.  TKs!
>>>> 
>>>> 73/jeff/ac0c
>>>> alpha-charlie-zero-charlie
>>>> www.ac0c.com
>>>> 
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[time-nuts] Re: Commercial solution - 122.88 Mhz low noise source

[Bob kb8tq]

Hi

Does the Abricon have an EFC input? If so, rigging up a PLL to lock 
122.88 to 10 MHz is probably the best option.

Bob

> On Apr 3, 2022, at 7:57 PM, Jeff Blaine  wrote:
> 
> I've got a couple of Red Pitaya 122-16 SDR and want to discipline them to an 
> external low noise GPSDO source.  Wanted to see if there were some easy 
> solutions that fell into this category.
> 
> For the prior generation (QS1R), I had built a homebrew OCXO based on a 
> custom low noise crystal from ICM and that worked great for the 125 Mhz 
> application.  Unfortunately this Pitaya is a bit removed in Fc and ICM has 
> been out of biz for several years now.
> 
> The RP's on-board oscillator is the respectable Abracon ABLNO 122.88 which 
> runs about -115 dBc/hz at 100 Hz spacing and that's probably not a limitation 
> for the unit.  Unfortunately the OSC is a bit too sensitive to external temp 
> and wanders around quite a lot (+/- 5 PPM) compared to the homebrew OCXO - 
> over the annual ambient temp range of (10-35C) (all datasheet referenced 
> values). The Abracon unit is mounted on the board with the FPGA and sees 
> additional temp range depending on other heat generation factors (FPGA 
> loading).
> 
> I have a couple of commercial GPSDO on the bench now, so a pretty clean 10 
> Mhz GPSDO reference is available now.  And I've seen enough board discussion 
> to know that a homebrew GPSDO solution is one of those "it's harder than it 
> looks" things.  Hence the desire just to buy whatever is the silver bullet, 
> as long as it does not require too much silver.  ha ha
> 
> Appreciate any suggestions.  TKs!
> 
> 73/jeff/ac0c
> alpha-charlie-zero-charlie
> www.ac0c.com
> 
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[time-nuts] Re: TIC reference clock in 3-cornered hat

[Bob kb8tq]

Hi

A lot depends on just what is being measured. 

If you are looking at three PPS signals, you *might* be running on 
the internal reference for a second. If you are comparing 10 MHz 
signals, the max time is 100 ns. Getting (say) 1 ps accuracy in the 
first means 1x10^-12 / 1 = 1 ppt on your local reference. In the 
second case it’s 1x10^12 / 100 x 10^-9 = 100,000 ( so 10 ppm). 
In the first case the local reference could matter a lot. In the second
case … not so much. 

The most common approach is to run the local references all off of 
a “good” standard and to align the signals so that it’s not a major 
issue in the results. ( so keep your 1 PPS signals close to each other).

Bob

> On Apr 3, 2022, at 5:50 PM, Marek Doršic  wrote:
> 
> Hi,
> 
> 3-cornered hat was mentioned several times past days. What reference 
> should be used for time interval counters comparing oscillator pairs?
> Should internal references be used what ever bad they are? A home standard 
> (4th oscillator)? The third oscillator C when comparing pair A-B? ...?
> 
> Thanks,
>.marek
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[time-nuts] Re: Low Phase Noise 10 MHz bench signal source sought

[Bob kb8tq]

Hi

These days a PLL is going to either be analog or digital. If it’s 
analog, you get into size constraints related to capacitors
as you go to lower crossover frequencies. With digital, you
get into all of the noise issues that any digital circuit will have.
(Yes, they can be addressed but it’s not easy at very low
offset frequencies). 

Regardless of design, you will always have some noise peaking.
> 89 degree phase margins can help with this, but they bring
in other problems. Setting the phase margin and other parameters
is either a mathematical design process or done with simulation. 
It can be very frustrating doing it by trial and error. I still find 
Phaselock Techniques by Floyd M Gardner to be a good reference 
on this stuff. 

The normal PLL control loops are fairly low order filters. If you
go high order, loop stability (and peaking) becomes difficult
to handle. Because it’s a simple filter, things like close in spurs 
will only be attenuated by some finite amount. 

So yes, you will have issues. Dealing with those issues means 
an area ( = range of offset frequencies) that are not as nice as you
might wish. That’s just the way the real world works. 

Bob

> On Apr 3, 2022, at 8:14 AM, ew via time-nuts  wrote:
> 
> There is a lot of talk of Rb's and OCXO's  and using an OCXO for clean up. 
> Very little about the clean up loop. It is key for overall performance. We 
> are working on it for quite some time and are not happy with the results. On 
> list off list would greatly be appreciated.   
>Bert Kehren  
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[time-nuts] Re: Can one use a two channel TC with an internal reference to measure two independent oscillators to establish the quality of the internal reference?

[Bob kb8tq]

Hi

Usually the best / quick / easy way to evaluate something like
a two channel counter is do it with a pair of signals that are independent
from each other, but very close to the same frequency. Rb’s are often
used, but the technique is not specific to them. 

You feed a PPS ( or whatever ) from each source into your channels. 
The reference for the counter is “something else”. As the two PPS
edges move along relative to each other, you sweep out the range
of the device. What you typically find is that the gizmo isn’t quite as
linear as you had hoped. You might even find dead spots ….

Bob

> On Apr 2, 2022, at 3:10 PM, Erik Kaashoek  wrote:
> 
> During evaluation of a two channel TC that can measure simultaneously the
> phase of two external oscillators the need came to understand the quality
> of the internal reference while eliminating (some) uncertainty of the
> performance of the two external oscillators.
> The 3 cornered hat method comes to mind as a way to get insight but I am
> sure I do not understand this all good enough.
> Some practical info on the setup:
> Two channel TC with 4 ns clock resolution and capable to measure phase
> against it's internal clock using interpolation over maximum 5 phase
> measurements per second.
> The external clocks being measured are a Rb with unknown performance and an
> OCXO with unknown performance.
> Any advice on how to create a better understanding of the performance of
> the internal reference using the above setup?
> Erik.
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[time-nuts] Re: Low Phase Noise70 10 MHz bench signal source sought

[Bob kb8tq]

Hi

Rb and low phase noise ( at least far removed ) are sort of mutually 
exclusive specs. You need to pick one … 

Assuming the decision is to go for the -170 dbc/Hz spec, Congratulations 
you are buying on OCXO. Not quite clear which OCXO, but it’s pretty likely
to be an OCXO. (Yes, there are exceptions, but they are rare enough to be 
in the “don’t bother” category), 

Next step would be to decide on the max offset that needs to cut in at. 
100 Hz is into the “crazy/ not gonna happen” region. 1KHz is unlikely. 
10 KHz is doable. 100 KHz relaxes things a bit. 

While the “buy a bunch and test” approach works for things like ADEV, 
it probably isn’t the best approach for this spec. If you buy a bunch of this
or that OCXO, their 10KHz phase noise *should* be pretty consistent. 
Sorting to get 2 db … nope, not worth it.  It *is* a pretty good bet that 
a commercial spec at 10K will be beat by 3 to 6 db. 

What to buy? Head off to the spec sheets on whatever you see on eBay
and make some guesses. 

Bob





> On Apr 1, 2022, at 5:12 PM, Joseph Gwinn  wrote:
> 
> I'm looking for suggestions for AC-powered 10 MHz sinewave laboratory 
> signal sources with very low phase noise, having a noise floor below 
> -170 dBc/Hz.  Rubidium is desired, but not essential.  Reliability 
> and durability in lab use is essential.
> 
> Which makes and models should I consider purchasing?  
> 
> I like the SRS model FS725, but its noise floor is too high at -150 
> dBm/Hz, 20 dB noisier than many things I may wish to measure.  
> 
> 
> Thanks,
> 
> Joe Gwinn
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[time-nuts] Re: 32.768Khz Crystal Trimming

[Bob kb8tq]

Hi

The typical 32 KHz crystal has a parabolic tempco. It peaks at kinda 
sorta room temperature. This makes it work pretty well on a normal 
wrist. 

As you get away from ~ 25C, the frequency drops. Since it’s a 
parabola, the further away from room you get the faster it drops. 
Something in the 1 to 2 ppm / C is not a bad guess for typical “on a 
board” sort of temperatures. 

It’s pretty hard to predict what temperature the board you just made 
will operate at out in the field. Thus there are limits on what you can 
expect from a watch crystal. 

10 seconds over 100,000 seconds ( so a bit over a day) is hundred
ppm. That’s a lot of shift and yes, it’s worth taking care of that
sort of offset. 

At 1 second per 100,000 you are at 10 ppm. That is in the range of 
“questionable temperature”. You can get it set that well ( or better) 
on your bench. As soon as it leaves your bench …. all that work may 
not have been worth the effort. 

No, temperature isn’t the only variable. The crystals have a calibration
tolerance at room ( +/- 20 ppm on this crystal). They most certainly do 
age ( 3 ppm  / year on this one). They also shift when you solder them 
into a board. That shift may or may not relax over the next few days. 

If your application *needs* a second a day sort of performance, 
something sort of improvement over a basic watch crystal will be 
needed. At the very least, the 20 ppm cal tolerance will need to be 
adressed. 

Bob

> On Apr 1, 2022, at 12:34 PM, Richard (Rick) Karlquist  
> wrote:
> 
> No one mentioned tempco, so I will.  Ideally you should do your
> calibration at a temperature corresponding to the long term
> average in your workshop.  If the crystal is in a piece of
> equipment with a temperate rise, it should be accounted for,
> and then going forward you have to leave the equipment powered
> up 24/7.  The crystal is probably a tuning fork, meaning it
> won't be AT cut.  It may have a substantial tempco around
> room temp.  In which case that old time-nuts insult may apply:
> 
> "congratulations, nice thermometer."
> 
> Rick N6RK
> 
> BTW, I go back 48 years with crystals.
> 
> On 4/1/2022 8:30 AM, Bernd Neubig wrote:
>> Hi,
>> If you do not want to make it a time-nuts style research project, but just
>> look for a quick fix - here is a rule of thumb:
>> This kind of crystal usually has a trimming sensitivity of around -10
>> ppm/pF. This means, if you increase the value of both capacitors on either
>> side by 2 pF will increase the load capacitance by 1 pF and thus lower the
>> frequency by  about 10 ppm.
>> If you need to vary one cap slightly more than the other, (to get a finer
>> resolution), please use the one at the output side of the on-chip oscillator
>> stage.
>> Take care of the start-up margin (safety to get a reliable start-up after
>> power on). The larger the capacitors, the lower becomes the margin,
>> sometimes the margin drops rather quickly. Therefore, after having made the
>> changes, test the start-up behavior by switching on and off several times,
>> preferably with a slow voltage ramp.
>> Enjoy the crystal world (as I did for the last 45 years and still doing).
>> Bernd
>> -Ursprüngliche Nachricht-
>> Von: Dan Kemppainen [mailto:d...@irtelemetrics.com]
>> Gesendet: Freitag, 1. April 2022 14:47
>> An: time-nuts@lists.febo.com
>> Betreff: [time-nuts] 32.768Khz Crystal Trimming
>> Hi,
>> I've got a 32.768Khz (USA number format) crystal on a RTCC oscillator of a
>> small micro, and it's running fast. Around 10 seconds per day or so.
>> This is a bit more than an order of magnitude more than the datasheet
>> states.
>> The 9 seconds per day error should be a good measurement. The RTCC is
>> running a 1 second counter, and that's being compared to a 1 second counter
>> derived by clocking the micro from a 10Mhz EXT clock reference.
>> This is consistent between multiple copies of the board.
>> I'm assuming, the C1/C2 load capacitors to ground needs to be higher in
>> value to trim that oscillator closer to the correct frequency. Is this
>> correct? Any quick back of the napkin calculations how much additional load
>> capacitance would be needed?
>> For Ref, this is the crystal:
>> https://abracon.com/Resonators/ABS06.pdf
>> ABS06-32.768KHZ-1-T
>> Thanks,
>> Dan
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